Endothelial cells line the luminal surface of the vascular bed and are thought to play an active role in the specific proteolytic breakdown of locally deposited fibrin, Todd, J. Pathol. Bacteriol., 78, 281 (1959); Astrup, in Process in Chemical Fibrinolysis and Thrombolysis, Davidson et al. eds., vol. 3, pp. 1-57, Raven Press, New York (1978). The potential of endothelium to initiate and control this process is emphasized by its capacity to synthesize and release plasminogen activators (PAs), Loskutoff et al., Proc. Natl. Acad. Sci. (USA), 74, 3903 (1977); Shepro et al., Thromb. Res., 18, 609 (1980); Moscatelli et al., Cell, 20, 343 (1980); Laug, Thromb. Haemostasis, 45, 219 (1981); Booyse et al., Thromb. Res., 24, 495 (1981), including both tissue-type and urokinase-type molecules, Levin et al., J. Cell Biol., 94, 631 (1982); Loskutoff et al., Blood, 62, 62 (1983). Endothelial cells can also produce inhibitors of fibrinolysis, Loskutoff et al., Proc. Natl. Acad. Sci. (USA), 74, 3903 (1977); Levin et al., Thromb. Res., 15, 869 (1979); Loskutoff et al., J. Biol. Chem. , 256, 4142 (1981); Dosne et al., Thromb. Res., 12, 377 (1978); Emeis et al., Biochem. Biophys. Res. Commun., 110, 392 (1983); Loskutoff et al., Proc. Natl. Acad. Sci. (USA), 80, 2956 (1983); Levin, Proc. Natl. Acad. Sci. (USA), 80, 6804 (1983).
Although these inhibitors probably serve important regulatory roles in controlling the fibrinolytic system of the vascular wall, little is known about their specificity, mode of action, or biochemical nature. The conclusion that these inhibitors are actually synthesized by endothelial cells is obscured somewhat by recent reports that cultured cells can bind and internalize protease inhibitors from serum-containing culture medium, Cohen, J. Clin. Invest., 52, 2793 (1973); Pastan et al., Cell, 12, 609 (1977); Rohrlich et al., J. Cell Physiol., 109, 1 (1981); McPherson et al., J. Biol. Chem., 256, 11330 (1981).
The possibility of producing relatively unlimited amounts of tissue-type plasminogen activator (t-PA) by recombinant DNA technology as described in British patent application GB 2,119,804 A, published Nov. 23, 1983, has generated much interest, both clinically and commercially. The conversion of the relatively inactive molecule into an extremely efficient thrombolytic agent by fibrin itself, suggests that t-PA can exist as an active enzyme only when localized to the fibrin-platelet thrombus itself. Thus, t-PA is considered to be a much more specific thrombolytic agent than urokinase-type plasminogen activator and streptokinase.
The interactions between t-PA and fibrin have raised the argument that natural inhibitors of t-PA are not necessary to regulate this system; i.e., regulation is achieved through the formation/dissolution of fibrin, and, thus, do not exist. It is clear that the existence of such inhibitors in human blood would complicate attempts to design a specific, efficient, and safe thrombolytic program based upon natural and genetically engineered t-PA. At the very least, calculations such as those of dose, treatment time and efficacy of treatment would be difficult to predict and/or monitor. This problem would be especially acute if inhibitor levels varied from individual to individual.
The existence of specific inhibitors of t-PA in plasma is a matter of some dispute, Collen, Thromb. Haemostas., 43, 77 (1980). In fact, it has been reported, Korninger et al., Thromb. Haemostas., 46, 662 (1981), that the activity of t-PA added to plasma had an in vitro half-life of 90 minutes as compared to an in vivo half-life of 2 minutes, Korninger et al., Thromb. Haemostas, 46, 658 (1981). Based upon these observations, those authors concluded that t-PA inhibition by plasma was physiologically unimportant.
That conclusion has recently been challenged in Kruithof et al., Prog. in Fibrinolysis, 6, 362 (1983). In Chmielewska et al., Thromb. Res., 31, 427 (1983), direct evidence was recently reported for the existence of a rapid inhibitor of t-PA in plasma. In all cases, this anti-t-PA activity was detected in the plasma of patients with or at risk to develop thrombotic problems; i.e., the very individuals most likely to receive t-PA therapy. This finding may account for the failure of Korninger et al., Thromb. Haemostas., 46, 662 (1981), to detect such an activity since they only examined the plasma of "normal" individuals. These reports on t-PA inhibitors represent little more than qualitative descriptions of an "activity" detected in the blood of some individuals.
Recently, an antifibrinolytic agent in cultured bovine endothelial cells was detected, Loskutoff et al., Proc. Natl, Acad. Sci. (USA), 80, 2956 (1983). This inhibitor is a major endothelial cell product and is an inhibitor of plasminogen activator since it san neutralize the activity of both fibrin-independent (urokinase-type) and fibrin-dependent (tissue-type) plasminogen activators (PAs). The observation that human platelets contain an immumologically similar inhibitor, Erickson et al., Haemostasis, 14 (1), 65 (1984) and J. Clin. Invest. 74, 1465 (1984), that is released by them in response to physiologically relevant stimuli, e.g., thrombin, and in parallel with other platelet proteins, e.g., Platelet Factor 4, emphasizes the potential importacute of this inhibitor in human biology. Antiserum to the plasminogen activator inhibitor (PAI) from bovine aortic endothelial cells (BAEs) has been employed to show that the human endothelial PAI as well as that from plasma, serum and platelets are related, i.e., immunologically similar. Erickson et al., Proc. Natl. Acad. Sci. USA, 82, 8710 (1985).
The inhibitor found by Loskutoff et al., Proc. Natl. Acad. Sci. (USA), 80, 2956 (1983), was purified from bovine aortic endothelial cell conditioned media by a combination of concanavalin A affinity chromatography and preparative sodium dodecyl sulfate-polyacrylamide gel electrophoresis (SDS-PAGE), and was shown to be a single chain glycoprotein of a molecular weight of 50,000 daltons, having an isoelectric point of 4.5-5 [van Mourik et al., J. Biol. Chem. 259, 14914 (1984)].
Recent evidence indicates that there are three immunologically distinct plasminogen activator inhibitors (PAIs). The first is that discussed above that is derived primarily from endothelial cells. The second, reported by Astedt et al., Thromb. Haemostasis, 53, 122 (1985) was isolated from placenta. The third, reported by Scott et al., J. Biol. Chem., 260 7029 (1985) is protease nexin.
The endothelial cell type PAI differs, in addition to immunologically, from placental PAI and protease nexin in that it inhibits both single chain and two chain tissue-type plasminogen activator (t-PA) as well as urokinase-type plasminogen activator (u-PA), while protease nexin and the placental PAI exhibit substantially no t-PA inhibition at physiological concentrations. Those latter two inhibitors do inhibit u-PA activity at physiological concentrations. Still further, endothelial PAI exhibits beta-mobility when analyzed by agarose zone electrophoresis while the other two PAIs do not. In addition the endothelial cell PAI is stable to low pH values (e.g. pH 3) and SDS (0.1%), while the other two inhibitors are rapidly inactivated by either of these treatments. [van Mourik et al., J. Biol. Chem., 259, 14914 (1984)].
The results discussed hereinafter illustrate that the endothelial cell type
exhibiting , beta-mobility is also present in human placental extracts as is the placenta PAi reported by Astedt et al., Thromb. Haemostasis, 53, 122 (1985). Since two types of PAI are obtainable from placenta, the human
hereinbefore referred to as of endothelial cell origin will usually be referred to as beta-PAI or endothelial PAI, or endothelial cell type PAI while the PAI first isolated from placenta is referred to as placental-type PAI or placental PAI.


The present invention relates generally to cloth spreading machines and in particular to a method and apparatus for spreading cloth in a uniform manner on a cutting surface.
Cloth spreading machines are widely used in the garment industry and are generally employed to deposit a layer or layers of cloth on a cutting surface. One type of prior art spreader is mounted above a stationary cutting table and is adapted to travel reciprocally above the table, dispensing a layer of cloth from a cloth supply spindle, as it traverses the length of the table. If multiple layers of cloth are to be laid, the spreader will either return to its starting position and dispense another layer of cloth on the table or alternately, a second layer will be deposited during the return travel of the spreader.
Many of the prior spreaders, include sensors for monitoring the tension in the cloth web. These sensors attempt to control the rate of cloth feed so that the cloth is spread onto the cutting surface uniformly at a predetermined tension, or "tension free".
In some prior art spreaders, a sensor in the form of a dancer roll is used to monitor cloth tension. The dancer roll is generally pivotally mounted and interposed in the web path so that changes in web tension will cause positional changes in the dancer roll. In one spreader, the dancer roll is operatively connected to a switch that controls the cloth drive; and in another spreader, the dancer roll is connected to a potentiometer which modifies the speed of the cloth drive.
Because most dancer rolls are spring biased, a predetermined tension in the web is necessary to overcome the dancer roll force and move it to an equilibrium position. In machines that use dancer rolls, it is difficult, if not impossible, to deposit a uniform layer of cloth on a cutting table, substantially "tension free", for the dancer roll itself will produce some tension in the web.
Other sensors have been suggested which do not contribute to the web tension. These include loop sensors which monitor the droop or sag in the web at a predetermined location. In one suggested spreader, the droop in the web is sensed optically whereas in another spreader, the droop is monitored by a radiant energy sensor. These sensing devices have not been totally satisfactory and in the case of the radiant energy sensor, have been rather expensive.
A variation in tension along the width of the cloth web is often encountered. Most prior art spreaders do not address this problem, but merely monitor the tension at one position on the web and adjust the cloth feed rate accordingly. Compensation for tension variation across the web is ignored. In the case of patterned fabric, specifically plaids, a variation in tension will skew the orientation of the cloth with respect to the cutting surface and a pattern mismatch will result when the garment is assembled.
With the advent of automatic cutting machines, new problems arise and the old problems are compounded. One type of automatic cutting machine is a laser cutter which includes a computer controlled laser head that directs a minute laser beam over a ply of cloth on a cutting surface and cuts out individual garment pieces while simultaneously fusing the edges. The cutting surface is a honey combed conveyor that extends to either side of the laser cutting station. A ply of cloth is laid upon the loading side of the conveyor surface and is then advanced into the cutting station. The cut pieces and the remnants are transported by the conveyor out of the cutting station to an unloading area.
Spreaders adapted for automatic cutting systems have been proposed. For the laser cutting system described above, a spreader has been proposed which is positioned at the input end of the conveyor and includes a cloth feed roll, an associated drive motor and an optical loop sensor for maintaining a predetermined tension in the web as it is deposited on the conveyor. In the suggested spreader, the cloth feed drive is controlled by the laser cutting machine and dispenses cloth onto the conveyor as it advances toward the cutting station. Because the conveyor speed is quite high, the cloth feed must accelerate quickly to match the speed of the conveyor. Any lag in the cloth feed mechanism will be manifested as an area of excessive tension or stretch in the cloth layer. Garment pieces cut from these stretched areas will not be dimensionally stable. The problem is further aggravated with pattern fabrics, especially plaids, for any nonuniformity is reflected in stripe or pattern skewing. In practice, it has been found that this spreader could not be made to spread cloth consistently on a high speed laser cutting system.
Another problem associated with this and other prior suggested spreaders is the inability of the operator adequately to inspect the cloth prior to advancing into the cutting station to insure that it has been spread uniformly. Moreover, even if the operator observes nonuniformity, no provision for rewinding the cloth onto the spreader is provided so that it can be re-spread.
Finally, it is quite common for remnants to remain on, and by adhered to, the cutting surface after leaving the unloading station. These remnants should be removed because the presence of one will cause distortions in subsequent spreading and cutting operations. Because the prior suggested spreaders are mounted at the input end of the conveyor, it is virtually impossible for the operator to inspect the conveyor prior to cloth spreading, and as a consequence, it is difficult to be certain that all remnants have been removed.


The present invention relates to wireless communication systems in general and, more particularly, to an improved method and apparatus for controlling the power of signals transmitted by base stations and mobile units operating in a CDMA communications system.
Many wireless communications systems in use today employ a form of spread-spectrum communications technology known as code-division multiple-access, or simply xe2x80x9cCDMAxe2x80x9d. In systems utilizing spread spectrum communications technology such as CDMA, signals transmitted by a base station or mobile unit are spread out over a very wide frequency range using pseudo-random noise sequences, making the signals relatively immune to frequency-dependent interference.
The majority of wireless communications systems using CDMA are based on TIA/EIA standard IS-95, which is incorporated by reference herein. IS-95 is known in the industry as a second generation standard. It is anticipated that there will soon be a new industry-wide CDMA standard emerging from various proposals currently under consideration by the International Telecommunications Union (ITU). Such proposals are collectively known as third generation (3G) CDMA proposals and include xe2x80x9ccdma2000 RTT Candidate Submission to ITU-Rxe2x80x9d, which is incorporated by reference herein.
In any CDMA system, the mobile units in a given cell act as geographically disparate signal sources which activate at random times. Consequently, it is not possible to synchronize reverse-link transmissions, i.e., transmission from the various mobile units to the cell""s base station. It is therefore impossible for a base station to perform accurate detection (even using pilot-assisted coherent reception as in 3G CDMA) without relying on a feedback mechanism to adjust the transmit power of each mobile unit. As a result, a dynamic method of power control known as closed-loop power control is commonly employed for controlling reverse-link power.
On the other hand, forward-link CDMA signals transmitted by a base station and destined for the various mobile units in the cell are designed to be mutually orthogonal and, furthermore, transmission of these signals can be synchronized by the base station. Thus, not only is a transmitted CDMA signal destined for a given mobile unit immune to interference from signals destined for other mobile units, but the mobile unit is able to perform coherent detection with a large processing gain. Accordingly, second generation standards such as IS-95 do not provide for closed-loop power control in the forward-link direction. Nevertheless, it has been found from experience with IS-95 CDMA that significant additional performance improvements can be achieved by using a feedback power control mechanism as in the reverse-link. Accordingly, most 3G CDMA proposals call for the use of closed-loop power control.
Closed-loop power control consists of a destination (which could be a base station or a mobile unit) measuring the signal-to-interference ratio of a signal received from a source and comparing the measured signal-to-interference ratio with a predetermined target value. If the measured value is greater than the target value, then the power transmitted by the source may be lowered, while if it is less than the target value, then the power transmitted by the source must be increased in order to meet the target. The desired power adjustment is forwarded by the destination back to the source along an appropriate power control subchannel.
A separate outer-loop power control mechanism is responsible for setting the target signal-to-interference ratio. The guidelines for implementing open-loop power control in the forward-link (3G CDMA) and the reverse-link (IS-95 and 3G CDMA) directions are fairly flexible. Still, thus far, designers have chosen to constrain the target signal-to-interference ratio to a constant value or one which depends on the type of service being offered. While this is satisfactory for situations where the data rate of a service is generally constant and quite low, many emerging applications require high data rates (certain 3G CDMA proposals anticipate that data connections at rates of up to 2 megabits per second per connection should be supported) and are characterized as having a substantially bursty traffic pattern.
As a result, if a conventional closed-loop power control algorithm relying on a fixed or service-specific target signal-to-interference ratio is applied in, say, the reverse-link direction, then whenever there is an increase in the instantaneous traffic from a given mobile unit, all the other mobile units will be required to raise their transmit power significantly because of interference from the given mobile unit. The fact that the other mobile units have raised their transmit power causes interference to the given mobile unit, which is again forced to increase its transmit power, and so on. This phenomenon will be repeated at a time scale of the burst duration which makes it a frequently occurring event.
Frequent increases in transmit power may cause undue stress on the transmitter power amplifier, which may consequently limit the cell capacity. Furthermore, the power control algorithm is required to converge quickly in order to avoid outages during the bit rate transition from a low bit rate to a high bit rate.
Similar effects occur in the forward-link direction, although other mobile units within the same cell as the given mobile unit are relatively immune to interference caused by forward-link transmissions involving the given mobile unit. Still, the above described effects can occur as a result of bursty forward-link transmissions occurring in neighbouring cells.
Clearly, the signal-to-interference ratio alone does not appear to be an adequate characterization of the grade of service required of a bursty link. For instance, when the mobile unit is transmitting at a high bit rate, its required performance and required quality of service are likely different from when the bit rate is considerably lower. Thus, there is a need in the industry to provide a method and apparatus for controlling transmitted power by a base station or mobile unit which takes into account the bursty nature of data traffic.
The invention can be summarized according to a first broad aspect as a method for use in a communications system wherein the transmit power of a wireless link is adjusted so that link performance meets a target level, the method including dynamically adjusting the target level as a function of the traffic characteristics of the link.
The invention can be summarized according to a second broad aspect as a method for use in a closed-loop power control system wherein the transmit power of a source unit communicating with a destination unit across a wireless link is varied in accordance with measured performance and a target performance parameter. The method is one of setting the target performance parameter and is performed at the destination unit. The method includes (a) detecting the start and end of data bursts received from the source unit across the link; and (b) if the performance of the link is adequate, then gradually increasing the target performance parameter when the start of a burst is detected and gradually decreasing the target performance parameter when the end of a burst is detected.
According to another broad aspect, the invention can be summarized as a method of generating power control commands for transmission to a source unit communicating with a destination unit. The method includes (a) measuring an instantaneous performance parameter of signals received from the source unit; (b) determining an instantaneous bit rate, denoted RMINST, of the signals received from the source unit; (c) computing a threshold instantaneous performance parameter as a first function of the measured instantaneous performance parameter, of RMINST, and of at least one target error performance-parameter; and (d) generating a power control command based upon a second function of the measured instantaneous performance parameter and the threshold instantaneous performance parameter.
Preferably, the measured instantaneous performance parameter is a measured instantaneous signal-to-interference ratio, denoted (S/I)MINST, and the threshold instantaneous performance parameter is a threshold instantaneous signal-to-interference ratio, denoted (S/I)LINST. Preferably, two of the target error performance parameters are a target average signal-to-interference ratio, denoted (S/I)*AVG, and a target maximum outage time, denoted T*MAX.
Preferably, step (c) includes (c1) integrating (S/I)MINST over a first time window to compute a measured average signal-to-interference ratio, denoted (S/I)MAVG; (c2) integrating RMINST over a second time window to compute an average bit rate, denoted RMAVG; and (c3) among a plurality of most recent values of (S/I)MINST, determining the longest amount of time during which (S/I)MINST was below (S/I)*AVG, said longest amount of time being denoted TMMAX.
Preferably, the first function is:                               (                      S            I                    )                INST        L            =                                    (                          S              I                        )                    AVG          *                ·        α              ,    where        α    =          {                                                  1              ,                                                                          if                                                  xe2x80x83                                                                  T                  MAX                  M                                             greater than                                                 T                  MAX                  *                                                                  xe2x80x83                                                or                                                  xe2x80x83                                                                                (                                          S                      I                                        )                                    AVG                  M                                             less than                                                 (                                      S                    I                                    )                                AVG                *                                                                                R                  AVG                  M                                                  R                  INST                  M                                            ,                                                          otherwise              .
The invention can be summarized according to another broad aspect as a computer-readable storage medium which, when processed by a computer at a destination unit, executes a sequence of steps to generate a power control command for transmission to a source unit communicating with the destination unit. The steps include (a) computing a threshold instantaneous performance parameter as a first function of a measured instantaneous performance parameter, a measured instantaneous bit rate and at least one target error performance parameter; and (b) generating a power control command based upon a second function of the measured instantaneous performance parameter and the threshold instantaneous performance parameter.
According to still another broad aspect, the invention may be summarized as a statistical power control block for use in a destination unit communicating with at least one source unit. The statistical power control block includes a unit for computing a threshold instantaneous performance parameter as a first function of a measured instantaneous performance parameter, a measured instantaneous bit rate and at least one target error performance parameter; and a unit connected to the computing means, for generating a power control command based upon a second function of the measured instantaneous performance parameter and the threshold instantaneous performance parameter.
The invention can also be summarized as a functional unit for generating power control commands. The functional unit includes a maximum selector for receiving a plurality of composite correlation levels and selecting the largest thereamong, as well as an integrator connected to the maximum selector, for receiving the largest composite correlation level and integrating it over a predetermined length of time, thereby to produce a measured instantaneous performance parameter at regular intervals. In addition, the functional unit includes the above-described statistical power control block.
When the invention is implemented, a sudden burst of data results in a lowering of the threshold instantaneous signal-to-interference ratio, until such time that the average signal-to-interference ratio or the maximum outage time are no longer acceptable. This provides a smoothing effect of the interference induced to other users and there may result an increase in cell capacity. Similarly, if the data rate is reduced while the limitations on the average error performance and outage times are being respected, then the average bit rate will be higher than the instantaneous bit rate, which increases the threshold instantaneous signal-to-interference ratio, which further maintains the mobile unit""s transmitted power at a relatively high power level for a longer time interval, followed by a gradual reduction as the average bit rate drops with time.


This invention relates generally to toy water shooters, and more particularly to a novelty toy water shooter that may be concealed under the user""s clothing.
Young people of all ages enjoy water fights with toy water guns. As such, it is not surprising there are many, many different types of water shooters available. These run from simple hand-held squirt guns that use trigger-activated pumps to eject water, to more complicated and sophisticated shooters that rely upon pressurized tanks to shoot a stream of water a significant distance.
A concealed water gun adds an extra dimension of fun to water fights. Among other advantages available when the water gun is hidden, the gun may go unnoticed until it is used to douse its target, and if the shooter is clever enough and the concealment good enough, even after the target has been hit. U.S. Pat. No. 4,997,110 discloses a prior concealed water shooter. While the gun disclosed in the ""110 patent may be concealed, it relies upon an electric pump powered by batteries and activated by an electric switch to eject water from the nozzle, and such electrical components add complexity to what is essentially a toy novelty.
There is a need therefore for improved toy water shooters, and in particular, concealable water guns.
A pressurizable bladder is configured for attachment to a user""s arm in a concealed position. The bladder has a refill tube for adding water, and an air inlet that is attached to a hand pump for pressurizing the bladder. A nozzle and valve are fluidly connected to the bladder and a trigger is operable to selectively open and close the valve.
In a second illustrated embodiment the pressure bladder is configured for attachment to the user""s waist, and the other components are modified accordingly. The hand pump and bladder-refill tube are connected to the bladder with conduitsxe2x80x94the hand pump conduit may be extended from the bladder down the user""s sleeve and held in one hand, and the nozzle and trigger are held in the user""s other hand.


1. Field of the Invention
The present invention relates to heregulin variants, nucleic acid molecules encoding such variants, and related vectors, host cells, pharmaceutical compositions, and methods. In particular, the invention relates to amino acid substitution variants of human heregulin-β1 having an enhanced affinity for the ErbB-3 and ErbB-4 receptors.
2. Description of the Related Art
Transduction of signals that regulate cell growth and differentiation is regulated in part by phosphorylation of various cellular proteins. Protein tyrosine kinases are enzymes that catalyze this process. Receptor protein tyrosine kinases are believed to direct cellular growth via ligand-stimulated tyrosine phosphorylation of intracellular proteins. Growth factor receptor protein tyrosine kinases of the class I subfamily include the 170 kilodalton (kDa) epidermal growth factor receptor (EGFR) encoded by the erbB1 gene. erbB1 has been causally implicated in human malignancy. In particular, increased expression of this gene has been observed in more aggressive carcinomas of the breast, bladder, lung, and stomach.
The second member of the class I subfamily, p185neu (also called the ErbB-2 receptor or p185HER2), was originally identified as the product of the transforming gene from neuroblastomas of chemically treated rats. The neu (erbB2 or HER2) gene encodes a 185 kDa receptor protein tyrosine kinase.
Amplification and/or overexpression of the human erbB2 gene correlates with a poor prognosis in breast and ovarian cancers. Slamon et al., Science 235:177-82 (1987); Slamon et al., Science 244:707-12 (1989). Overexpression of erbB2 has been correlated with other carcinomas including carcinomas of the stomach, endometrium, salivary gland, lung, kidney, colon and bladder. Accordingly, in U.S. Pat. No. 4,968,603, Slamon et al. describe and claim various diagnostic assays for determining erbB2 gene amplification or expression in tumor cells. Slamon et al. discovered that the presence of multiple copies of the erbB2 oncogene in tumor cells indicates that the disease is more likely to spread beyond the primary tumor site, and that the disease may therefore require more aggressive treatment than might otherwise be indicated by other diagnostic factors. Slamon et al. conclude that the erbB2 gene amplification test, together with the determination of lymph node status, provides greatly improved prognostic utility.
A further related gene, called erbB3 (or HER3), which encodes the ErbB-3 receptor (p180HER3) has also been described. See U.S. Pat. No. 5,183,884; Kraus et al., PNAS USA 86:9193-97 (1989); EP Patent Application No. 444,961A1; Kraus et al., PNAS USA 90:2900-04 (1993). Kraus et al. (1989) discovered that markedly elevated levels of erbB3 mRNA were present in certain human mammary tumor cell lines indicating that erbB3, like erbB1 and erbB2, may play a role in human malignancies. Also, Kraus et al. (1993) showed that EGF-dependent activation of the ErbB-3 catalytic domain of a chimeric EGFR/ErbB-3 receptor resulted in a proliferative response in transfected NIH-3T3 cells. Furthermore, these researchers demonstrated that some human mammary tumor cell lines display a significant elevation of steady-state ErbB-3 receptor tyrosine phosphorylation, further implicating this receptor in human malignancies. The role of erbB3 in cancer has been explored by others, and this gene has been found to be overexpressed in breast (Lemoine et al., Br. J. Cancer 66:1116-21 [1992]), gastrointestinal (Poller et al., J. Pathol. 168:275-80 [1992]; Rajkumer et al., J. Pathol. 170:371-78 [1993]; Sanidas et al., Int. J. Cancer 54:935-40 [1993]), and pancreatic cancers (Lemoine et al., J. Pathol. 168:269-73 [1992], and Friess et al., Clinical Cancer Research 1:1413-20 [1995]).
The class I subfamily of growth factor receptor protein tyrosine kinases has been further extended to include the ErbB-4 (HER4) receptor, which is the product of the erbB4 (HER4) gene. See EP Patent Application No. 599,274; Plowman et al., PNAS USA 90:1746-50 (1993); and Plowman et al., Nature 366:473-75 (1993). Plowman et al. found that increased erbB4 expression closely correlated with certain carcinomas of epithelial origin, including breast adenocarcinomas. Diagnostic methods for detection of human neoplastic conditions (especially breast cancers) that evaluate erbB4 expression are described in EP Patent Application No. 599,274.
The quest for the activator of the erbB2 oncogene has lead to the discovery of a family of heregulin polypeptides. In humans, the heregulin polypeptides characterized thus far are derived from alternate splicing of a single gene which was mapped to the short arm of chromosome 8 by Lee and Wood, Genomics 16:790-91 (1993).
Holmes et al. isolated and cloned a family of polypeptide activators for the ErbB-2 receptor which they called heregulin-α (HRG-α), heregulin-β1 (HRG-β1), heregulin-β2 (HRG-β2), and heregulin-β3 (HRG-β3). See Holmes et al., Science 256:1205-10 (1992); WO 92/20798; and U.S. Pat. No. 5,367,060. These researchers demonstrated the ability of the purified heregulin polypeptides to activate tyrosine phosphorylation of the ErbB-2 receptor in MCF7 breast tumor cells. Furthermore, the mitogenic activity of the heregulin polypeptides on SK-ER-3 cells (which express high levels of the ErbB-2 receptor) was also demonstrated.
Heregulins are large multi-domain proteins that are typically expressed as “pro-heregulins.” Pro-heregulins have been shown to undergo proteolytic processing to a mature soluble form (usually of about 44-45 kDa). Processing has been shown to occur intracellularly or at the cell surface. Domains in the soluble form include (in order from the N- to the C-terminus) an immunoglobulin homology (Ig-like) domain, a spacer region rich in glycosylation sites, and a domain similar to a domain found in EGF that is sufficient for ErbB receptor binding and activation. See Barbacci, et al., J. Biol. Chem. 270:9585-89 (1995).
The heregulin EGF-like domains are characterized by substantial structural similarities to (Jacobsen et al., Biochemistry 35:3402-17 [1996]), and limited sequence homology with, EGF residues 1-48 (Holmes, et al., supra). Functional similarities between the heregulin EGF-like domains and EGF have been established by data showing that blocks of EGF sequence substituted into heregulin-β1 do not impair binding to cells co-expressing ErbB-3 and ErbB-2. Barbacci et al., supra.
While heregulins are substantially identical in the first 213 amino acid residues, they are classified into two major types, α and β, based on two EGF-like domains that differ in their C-terminal portions. For example, the heregulin-α EGF-like domain differs from that of the β1-isoform by nine substitutions near the C-terminus. The β-isoform has been reported to bind ErbB receptors with approximately eight to 10-fold higher affinity than the α-isoform. Wen et al., Mol. Cell. Biol. 14:1909-19 (1994).
The solution structure of the heregulin-α EGF domain has recently been determined at high resolution by NMR: Jacobsen et al., supra; Nagata et al., EMBO J. 10, 3517-3523 (1994). The salient features of this domain include (1) an N-terminal subdomain containing a central three-stranded β-sheet with an intermittent helix and (2) a smaller C-terminal subdomain that contains a short stretch of β-sheet. The EGF domain is stabilized by three disulfide bonds, two in the N-terminal subdomain and one the C-terminal subdomain. The pairing of the six corresponding cysteine residues is conserved in EGF-like domains from all heregulins and from EGF.
The 44 kDa neu differentiation factor (NDF), which is the rat equivalent of human HRG, was first described by Peles et al., Cell, 69:205-16 (1992), and Wen et al., Cell, 69:559-72 (1992). Like the human heregulin polypeptides, NDF has an Ig-like domain followed by an EGF-like domain and lacks a N-terminal signal peptide. Subsequently, Wen et al. carried out “exhaustive cloning” to extend the family of NDFs. Wen et al., Mol. Cell. Biol., 14:1909-19 (1994). This work revealed six distinct fibroblastic pro-NDFs. Adopting the nomenclature of Holmes et al., the NDFs were classified as either α or β polypeptides based on the sequences of the EGF-like domains. Isoforms 1 to 4 are characterized on the basis of a variable region between the EGF-like domain and transmembrane domain. Also, isoforms a, b and c are defined based on variable-length cytoplasmic domains. These researchers conclude that different NDF isoforms are generated by alternative splicing and perform distinct tissue-specific functions. See also EP 505 148; WO 93/22424; and WO 94/28133 (discussing NDF).
Falls et al., Cell 72:801-815 (1993) describe another member of the heregulin family which they call “acetylcholine receptor inducing activity (ARIA) polypeptide.” The chicken-derived ARIA polypeptide stimulates synthesis of muscle acetylcholine receptors. See WO 94/08007. ARIA is a β-type heregulin and lacks the entire spacer region between the Ig-like domain and EGF-like domain of HRG-α and HRGβ1-β3.
Marchionni et al., Nature 362:312-318 (1993) identified several bovine-derived proteins that they call “glial growth factors (GGFs).” These GGFs share the Ig-like domain and EGF-like domain with the other heregulin proteins described above, but also have an amino-terminal kringle domain. GGFs generally do not have the complete spacer region between the Ig-like domain and EGF-like domain. Only one of the GGFs, GGFII, has an N-terminal signal peptide. See also WO 92/18627; WO 94/00140; WO 94/04560; WO 94/26298; WO 95/32724 (describing GGFs and uses thereof).
Ho et al. describe another member of the heregulin family called “sensory and motor neuron-derived factor (SMDF).” Ho et al., J. Biol. Chem. 270:14523-32 (1995). This protein has an EGF-like domain characteristic of all other heregulin polypeptides but a distinct N-terminal domain. In addition, SMDF lacks both the Ig-like domain and the spacer region found in other heregulin polypeptides. Another feature of SMDF is the presence of two stretches of hydrophobic amino acids near the N-terminus.
While the heregulin polypeptides were first identified based on their ability to activate the ErbB-2 receptor (see Holmes et al., supra), it has been discovered that certain ovarian cells expressing neu (erbB2) and neu-transfected fibroblasts did not bind or crosslink to NDF, nor did they undergo tyrosine phosphorylation in response to NDF. Peles et al., EMBO J. 12:961-71 (1993). This finding indicated that another cellular component was necessary for conferring full heregulin responsiveness.
Carraway et al. subsequently demonstrated that 125I-rHRG-β1 177-244 bound to NIH-3T3 fibroblasts stably transfected with bovine erbB3 but not to non-transfected parental cells. These researchers also expressed bovine ErbB-3 receptor in insect cells and showed that HRG-β1 177-244 bound to a preparation of ErbB-3 receptor solubilized from these cells. They concluded that ErbB-3 is a receptor for heregulin and mediates phosphorylation of intrinsic tyrosine residues as well as phosphorylation of ErbB-2 receptor in cells that express both receptors. Carraway et al., J. Biol. Chem. 269:14303-05 (1994). Sliwkowski et al. found that cells transfected with erbB3 alone show low affinities for heregulin, whereas cells transfected with both erbB2 and erbB3 show higher affinities. Sliwkowski et al., J. Biol. Chem. 269: 14661-65 (1994).
Plowman and his colleagues have similarly studied ErbB-4/ErbB-2 receptor activation. They expressed the ErbB2 receptor alone, the ErbB4 receptor alone, or the two receptors together in human T lymphocytes and demonstrated that heregulin is capable of stimulating tyrosine phosphorylation of ErbB-4, but could only stimulate ErbB-2 phosphorylation in cells expressing both receptors. Plowman et al., Nature 336:473-75 (1993).
These observations are consistent with the “receptor cross-talking” concept described previously by Kokai et al., Cell 58:287-92 (1989), Stern et al., EMBO J. 7:995-1001 (1988), and King et al., 4:13-18 (1989). These researchers found that binding of EGF to the EGFR resulted in activation of the EGFR kinase domain and cross-phosphorylation of the ErbB-2 receptor. This is believed to be a result of ligand-induced receptor heterodimerization and the concomitant cross-phosphorylation of the receptors within the heterodimer. Wada et al., Cell 61:1339-47 (1990).
Thus, the ErbB receptors are believed to be activated by ligand-induced receptor dimerization. Specifically, heregulins can bind separately to ErbB-3 and ErbB-4 receptors, but not to the ErbB-2 receptor. However, ErbB-2 is required for signaling, and heterodimers containing ErbB-2 in combination with ErbB-3 or ErbB-4 bind heregulins with higher affinity than homodimers containing ErbB-3 or ErbB-4. Plowman et al., Nature 366:473-75 (1993); Sliwkowski et al., J. Biol. Chem. 269:14661-65 (1994).
The biological activities of heregulins have been investigated by several groups. For example, Holmes et al. (supra) found that heregulin exerts a mitogenic effect on mammary cell lines (such as SK-SR-3 and MCF-7). Lewis et al. reported that heregulin-β1 stimulated proliferation and enhanced colony formation in soft agar in a number of human breast and ovarian tumor cell lines. Lewis et al., Cancer Research 56:1457-65 (1996). These researchers also showed that ErbB-2 is a critical mediator of heregulin responsiveness.
Pinkas-Kramarski et al. found that NDF (rat heregulin) is expressed in neurons and glial cells in embryonic and adult rat brain and primary cultures of rat brain cells, and suggested that NDF may act as a survival and maturation factor for astrocytes. Pinkas-Kramarski et al., PNAS USA 91:9387-91 (1994). Danilenko et al. reported that the interaction of NDF and the ErbB-2 receptor is important in directing epidermal migration and differentiation during wound repair. Danilenko et al., Abstract 3101, FASEB 8 (4-5):A535 (1994).
Meyer and Birchmeier analyzed expression of mouse heregulin during embryogenesis and in the perinatal animal using in situ hybridization and RNase protection experiments. Meyer and Birchmeier, PNAS USA 91:1064-68 (1994). These authors conclude, based on expression of this molecule, that heregulin plays a role in vivo as a mesenchymal and neuronal factor. Their findings also indicated that heregulin functions in the development of epithelia.
Falls et al. (supra) found that chicken ARIA plays a role in myotube differentiation, namely affecting the synthesis and concentration of neurotransmitter receptors in the postsynaptic muscle cells of motor neurons. Corfas and Fischbach demonstrated that ARIA also increases the number of sodium channels in chick muscle. Corfas and Fischbach, J. Neuroscience 13:2118-25 (1993).
Bovine GGFs have been reported to be mitogenic for Schwann cells. See, e.g., Brockes et al., J. Biol. Chem. 255:8374-77 (1980); Lemke and Brockes, J. Neurosci. 4:75-83 (1984); Brockes et al., J. Neuroscience 4:75-83 (1984); Brockes et al., Ann. Neurol. 20:317-22 (1986); Brockes, Methods in Enzym. 147:217-225 (1987); Marchionni et al., supra. Schwann cells provide myelin sheathing around the axons of myelinated neurons and thus play an important role in the development, function and regeneration of peripheral nerves. The implications of this role from a therapeutic standpoint have been addressed by Levi et al., J. Neuroscience 14:1309-19 (1994). Levi et al. discussed the potential for construction of a cellular prosthesis including Schwann cells that could be transplanted into areas of damaged spinal cord. Methods for culturing Schwann cells ex vivo have been described. See WO 94/00140; Li et al., J. Neuroscience 16:2012-19 (1996).
GGFII has been shown to be mitogenic for subconfluent quiescent human myoblasts, and differentiation of clonal human myoblasts in the continuous presence of GGFII results in greater numbers of myotubes after six days of differentiation. Sklar et al., J. Cell Biochem., Abst. W462, 18D, 540 (1994); see also WO 94/26298.
The relationship between the structure and function of new proteins can be investigated using any of a variety of available mutational analysis techniques. Examples of such techniques include alanine scanning mutagenesis and phagemid display. Alanine scanning can be used to identify active residues (i.e., residues that have a significant effect on protein function) in a protein or protein domain. For example, Cunningham and Wells used alanine scanning to identify residues in human growth hormone that were important for binding its receptor. Cunningham and Wells, Science 244:1081-85 (1999). In alanine scanning, a gene encoding the protein or domain to be canned is inserted into an expression vector, and mutagenesis is carried out to generate a series of vectors that encode proteins or domains in which sequential residues are converted to alanine. The encoded proteins or domain are expressed from these vectors, and the activities of the alanine-substituted variants are then tested to identify those with altered activity. An alteration in activity indicates that the residue at the alanine-substituted position is an active residue.
Phagemid display was developed to allow the screening of a large number of variant polypeptides for a particular binding activity. Smith and Parmley demonstrated that foreign peptides can be “displayed” efficiently on the surface of filamentous phage by inserting short gene fragments into gene III of the fd phage. Smith, Science 228:1315-17 (1985); Parmley and Smith, Gene 73:305-18 (1985). The gene III coat protein is present in about five copies at one end of the phage particle. The modified phage were termed “fusion phage” because they displayed the foreign peptides fused to the gene III coat protein. As each fusion phage particle displayed approximately five copies of the fusion protein, this mode of phage display was termed “polyvalent display.”
Scott et al. and Cwirla et al. showed that fusion phage libraries could be screened by sequential affinity selections known as “panning.” Scott et al., Science 249:386-90 (1990); Cwirla et al., PNAS USA 87:6378-82 (1990). However, early efforts to select high affinity fusion phage failed, presumably due to the polyvalence of the phage particles. This problem was solved with the development of a “monovalent” phage display system in which the fusion protein is expressed at a low level from a phagemid and a helper phage provides a large excess of wild-type coat protein. Bass et al., Proteins 8:309-14 (1990); Lowman et al., Biochem. 30:10832-38 (1991). Monovalent phage display can be used to generate and screen a large number of variant polypeptides to isolate those that bind with high affinity to a target of interest.


The present invention relates generally to the field of electrical control and monitoring systems, and more particularly to a system and method that integrates functions control and monitoring into a human machine interface.
A wide range of systems are known and are currently in use for controlling and monitoring processes, particularly in the industrial context. Such processes may generally include a large number of components, such as pumps, valves, conveyors, material handling and machining systems, and so forth. In most applications a significant number of the components are operated by prime movers such as electric motors. These may be manually operated and inspected, but are often more effectively controlled through programmed systems including protective components as well as activating and control components.
One type of industrial control center that has been developed over recent years in generally referred to as a motor control center. Such systems may provide for highly integrated control of a large number of devices and can be equipped for remote control and reporting functions. Moreover, such remote control, typically via industrial control networks, is often highly desirable because it permits locating the protective and power control components roughly in the vicinity of the controlled equipment, which may be quite distant from the more centralized control or monitoring facility or room where human operators are based.
Challenges faced in implementing and maintaining complex control systems include the configuration of the control system and the representation of the system in a manner that can be quickly and easily mastered by human operators. Moreover, during operation, the representations of the system provided to the human operators, and the meaning of the various controls at the operators' disposal are of considerable importance insomuch as they permit the operator to make efficient and timely decisions based upon accurate understanding of the current and future conditions of the system. Planning, programming and configuration of such systems and networks is, however, quite difficult.
In addition to the programming concerns, operating the modem industrial control and monitoring systems often involves different programs that are utilized by the system. These separate programs each require extensive programming to provide information that allows the operator to monitor and control various processes. For instance, one program may include detailed information relating to individual programmable devices, such as manuals and drawings, while another program may provide functionality that enables an operator to view more the complete system that is operating the processes.
There is a need, therefore, for an improved technique which may be employed to more effectively present and access information relating to an overall process or system in a human machine interface, while providing detailed and current information on specific components of interest upon demand by an operator. There is a particular need for a technique which will provide greater uniformity and consistency for the operation of the processes by integrating the information of various levels within the modern industrial control and monitoring systems, reducing development and programming time associated with the various programs, and enabling the operator to perform a greater level of diagnostics from a single interface.


Manufactured products often contain orifices and cavities or other hollow parts that result from the manufacturing process and/or that are designed into the product for various purposes, such as weight reduction. Automotive vehicles, for example, include several such orifices and cavities throughout the vehicle, including in the vehicle's structural pillars and in the sheet metal of the vehicle doors. It is often desirable to seal such orifices and cavities so as to minimise noise, vibrations, fumes, dirt, water and the like from passing from one area to another within the vehicle by means of sealing members or baffle elements built into the orifice or cavity. Likewise, such members or elements often fulfill an additional task of reinforcing the hollow structure of the manufactured product, e.g. automotive part, so much that it becomes more resistant to mechanical stress but still maintains the low weight advantage of the hollow structure.
Such elements used for sealing, baffling or reinforcing often consist of a carrier, made of plastic, metal, or another rigid material, and one or more layers of a thermoplastic material attached to it which is able to expand its volume when heat or another physical or chemical form of energy is applied. With such a design, it is possible to insert the baffle or reinforcement element into the hollow part of the structure during the manufacture process but also to leave the inner walls of the structure still accessible (or the cavities passable) by e.g. a liquid. For example, during the manufacture process of a vehicle, the hollow parts of a metal frame can still be largely covered by an electrocoating liquid while the baffle or reinforcement elements are already inserted, and afterwards during a heat treatment step, the expandable thermoplastic material layers of the baffle or reinforcement expand to close the cavities as intended.
The development of such baffles or reinforcement elements has led to highly advanced systems, where the expandable material is able to increase its volume by up to 2000% or more, forming a foam-like structure filling the cavities and adhering to the walls of the structure intended to be sealed, baffled, or reinforced. Especially in automotive manufacturing, this has led to considerable weight reduction and excellent dampening of noise or vibrations in the car body.
With advanced materials that are able to expand even more, it is possible to reduce initial material mass and therefore contribute further to weight reduction and cost efficiency.
However, increasingly small portions of expandable material can also cause problems in manufacturing of the baffle or reinforcement elements. As such elements (or at least the expandable material layers) are normally produced by injection molding or extrusion, the feeding of very small sections gives rise to higher processing demands, such as higher injection pressures, or causes quality issues, such as flashing, short shots (incompletely molded parts) or material degradation.
It is thus desirable to obtain a way of manufacturing baffle or reinforcement elements with less initial mass of expandable material, but without the problems connected to very small initial volumes of that material.


1. Field of the Invention
The present invention generally relates to executing instructions in a processor. Specifically, this application is related to increasing the efficiency of a processor executing branch instructions.
2. Description of the Related Art
Modern computer systems typically contain several integrated circuits (ICs), including a processor which may be used to process information in the computer system. The data processed by a processor may include computer instructions which are executed by the processor as well as data which is manipulated by the processor using the computer instructions. The computer instructions and data are typically stored in a main memory in the computer system.
Processors typically process instructions by executing the instruction in a series of small steps. In some cases, to increase the number of instructions being processed by the processor (and therefore increase the speed of the processor), the processor may be pipelined. Pipelining refers to providing separate stages in a processor where each stage performs one or more of the small steps necessary to execute an instruction. In some cases, the pipeline (in addition to other circuitry) may be placed in a portion of the processor referred to as the processor core. Some processors may have multiple processor cores, and in some cases, each processor core may have multiple pipelines. Where a processor core has multiple pipelines, groups of instructions (referred to as issue groups) may be issued to the multiple pipelines in parallel and executed by each of the pipelines in parallel.
As an example of executing instructions in a pipeline, when a first instruction is received, a first pipeline stage may process a small part of the instruction. When the first pipeline stage has finished processing the small part of the instruction, a second pipeline stage may begin processing another small part of the first instruction while the first pipeline stage receives and begins processing a small part of a second instruction. Thus, the processor may process two or more instructions at the same time (in parallel).
Processors typically provide conditional branch instructions which allow a computer program to branch from one instruction to a target instruction (thereby skipping intermediate instructions, if any) if a condition is satisfied. If the condition is not satisfied, the next instruction after the branch instruction may be executed without branching to the target instruction. Typically, the outcome of the condition being tested is not known until the conditional branch instruction is executed and the condition is tested. Thus, the next instruction to be executed after the conditional branch instruction may not be known until the branch condition is tested.
Where a pipeline is utilized to execute instructions, the outcome of the conditional branch instruction may not be known until the conditional branch instruction has passed through several stages of the pipeline. Thus, the next instruction to be executed after the conditional branch instruction may not be known until the conditional branch instruction has passed through the stages necessary to determine the outcome of the branch condition. In some cases, execution of instructions in the pipeline may be stalled (e.g., the stages of the pipeline preceding the branch instruction may not be used to execute instructions) until the branch condition is tested and the next instruction to be executed is known. However, where the pipeline is stalled, the pipeline is not being used to execute as many instructions in parallel (because some stages before the conditional branch are not executing instructions), causing the benefit of the pipeline to be reduced and decreasing overall processor efficiency.
In some cases, to improve processor efficiency, branch prediction may be used to predict the outcome of conditional branch instructions. For example, when a conditional branch instruction is encountered, the processor may predict which instruction will be executed after the outcome of the branch condition is known. Then, instead of stalling the pipeline when the conditional branch instruction is issued, the processor may continue issuing instructions beginning with the predicted next instruction.
However, in some cases, the branch prediction may be incorrect (e.g., the processor may predict one outcome of the conditional branch instruction, but when the conditional branch instruction is executed, the opposite outcome may result). Where the outcome of the conditional branch instruction is mispredicted, the predicted instructions issued subsequently to the pipeline after the conditional branch instruction may be removed from the pipeline and the effects of the instructions may be undone (referred to as flushing the pipeline). Then, after the pipeline is flushed, the correct next instruction for the conditional branch instruction may be issued to the pipeline and execution of the instructions may continue. Where the outcome of a conditional branch instruction is incorrectly predicted and the incorrectly predicted group of instructions is flushed from the pipeline, thereby undoing previous work done by the pipeline, the efficiency of the processor may suffer.
Accordingly, what is needed is an improved method and apparatus for executing conditional branch instructions and performing branch prediction.


The present invention relates in general to a bag packing apparatus, and more particularly, an apparatus for automatically opening, packing and sealing, successively, conveyed paper or plastic bags or the like.
In a heretofore known bag packing apparatus as shown in FIG. 1, wherein three-sided sealed bags 1, made of paper or plastic films with mouths 2 along their top edge, are charged with contents by means of a charging device 3 and tightly sealed 4, it was a common practice to convey the bags 1 successively in the direction of their width by means of chain conveyors 5. However, such apparatuses in the prior art have the following shortcomings, and consequently, many problems remain unsolved for establishing common availability for different bag sizes and for speeding up processing capability. For example, when the sidth of the bags changed it was necessary to change the mounting distance in the widthwise direction between the bag gripping claws 6 within the conveyor pitch P.sub.1 for each attachment. This resulted in a very complex structure and adjustment requirements for the machine; size change was not easy. In addition, in order to enhance the processing capability, the conveying speed had to be raised to a higher speed due to the fact that the bags were conveyed in the widthwise direction, and especially when the contents of the bags were liquid, there was a disadvantage in that the liquid would squeeze out of the top edge of the bag due to acceleration, resulting in further adverse effects upon sealing.


Electroacoustic transducers are capable of converting electric energy to acoustic energy and vice versa. Electroacoustic receivers typically convert electric energy to acoustic energy through a motor assembly having a movable armature. Typically, the armature has one end that is free to move while the other end is fixed to a housing of the receiver. The assembly also includes a drive coil and one or more magnets, both capable of magnetically interacting with the armature. The armature is typically connected to a diaphragm near its movable end. When the drive coil is excited by an electrical signal, it magnetizes the armature. Interaction of the magnetized armature and the magnetic fields of the magnets causes the movable end of the armature to vibrate. Movement of the diaphragm connected to the armature produces sound for output to the human ear. Examples of such transducers are disclosed in U.S. Pat. Nos. 3,588,383, 4,272,654 and 5,193,116.
The sound pressure output of a receiver is created by the travel, or deflection, of the armature when it vibrates. Maximum deflection of the moving armature creates maximum sound pressure output for a given armature geometry. The maximum deflection of an armature is limited by the magnetic saturation of the armature, which is governed by the maximum magnetic flux that the armature geometry can allow to pass therethrough. Therefore, the magnetic flux must be increased in order to increase the sound pressure output. The magnetic flux is limited by material type and cross-sectional area of the armature. Although an increase in the cross-sectional area causes a proportional increase in the maximum magnetic flux, the relative stiffness of the armature increases as well. Thus, merely increasing the cross-sectional area of the armature geometry does not provide a significant improvement in the maximum deflection of the armature.
The present invention addresses these and other problems.


Nearly one in seven people in the United States suffer from some type of chronic sleep disorder, and only 50% of people are estimated to get the recommended seven to eight hours of sleep each night. It is further estimated that sleep deprivation and its associated medical and social costs (loss of productivity, industrial accidents, etc.) exceed $150 billion per year. Excessive sleepiness can deteriorate the quality of life and is a major cause of morbidity and mortality due to its role in industrial and transportation accidents. Sleepiness further has undesirable effects on motor vehicle driving, employment, higher earning and job promotion opportunities, education, recreation, and personal life.
Primary sleep disorders affect approximately 50 million Americans of all ages and include narcolepsy, restless legs/periodic leg movement, insomnia, and most commonly, sleep apnea. Sleep apnea is defined as the cessation of breathing during sleep. The three major types of sleep apnea are obstructive sleep apnea (OSA), central sleep apnea (CSA), and complex sleep apnea (CompSA). Of these three, CSA is rare, while OSA is the most common. CompSA is a relatively new disease state that manifests itself after therapy is applied. Patients with CompSA are characterized by the emergence of new CSA events after the application of Continuous Positive Airway Pressure (CPAP). OSA's prevalence in society is comparable with diabetes, asthma, and the lifetime risk of colon cancer. OSA is grossly under diagnosed; an estimated 80-90% of persons afflicted have not received a clinical diagnosis. OSA is characterized by repetitive pauses in breathing during sleep due to the obstruction and/or collapse of the upper airway (throat), usually accompanied by a reduction in blood oxygen saturation, and often followed by an awakening to breathe (an apnea event). Respiratory effort continues during the episodes of OSA. Multiple episodes of apnea may occur in one night, causing sleep disruption. CSA is a neurological condition causing cessation of all respiratory effort during sleep, usually with corresponding decreases in blood oxygen saturation. In contrast to OSA, where there is respiratory effort from the brain stem but a physical blockage prevents inhalation of oxygen, in CSA the brainstem center controlling breathing shuts down, resulting in no respiratory effort and no breathing. The subject is aroused from sleep by an automatic breathing reflex. Frequent activation of the reflex results in very little sleep for the subject. The neurological mechanism behind CSA is very different from the physical cause of OSA. Although the effects of CSA and OSA are highly similar, effective treatment can differ. CompSA can be thought of as a combination of OSA and CSA. As mentioned before, CompSA is characterized by an emergence of CSA events after CPAP initiation.
Medications, hygiene, or some physical form of therapy can be used to treat apneas. Treatment of OSA and CSA vary substantially, which makes a proper diagnosis of the correct type of sleep apnea (OSA, CSA, or CompSA) critical for an effective treatment. Apnea treatment is provided based on the type of apnea, and can be adjusted by re-testing the subject at some later time to determine whether the condition or the symptoms have been alleviated. The most common method of treating OSA is continuous positive airway pressure (CPAP) and positive airway pressure (PAP) devices applied to the subject's airway to force the subject to breathe. When using a simple CPAP device to treat OSA, the air pressure acts as a splint, holding the airway open and reducing or removing the obstruction. The optimal pressure is determined by a sleep technician during a single titration night. The sleep technician manually adjusts the device to deliver a minimum pressure sufficient to force the airway open and reduce the number of apneas. Once the optimal pressure is determined, the device is programmed to consistently provide this pressure, and the patient is sent home.
Slightly more advanced PAP devices automatically adjust the air pressure based on sensors built into the device. The sensors measure gas flow, pressure, or other fluid characteristics in the device or its mask, and adjust the delivered pressure based on various algorithms known in the art. These auto-PAP devices rely on the single physiological variable (the measured or estimated fluid characteristics) to predict or detect an apnea event.
None of the devices on the market can be used to adjust the gas flow delivered to a subject based on a comprehensive evaluation of the subject's current physiological state or the subject's current symptoms. Further, none of the current devices can use a rich data set to predict or detect apnea and provide appropriate treatment. Still further, none of the current CPAP or PAP titration methods use a rich set of data over single or multiple nights to set the optimal pressure and other parameters. Still even further, none of the current devices can be used to automatically adjust a treatment device based on the subject's physiological signals. Still even further, none of the current titration devices can be used in the subject's home.
It is therefore an object of the present invention to adjust the treatment gas flow or pressure delivered to the subject based on the subject's current physiological state or symptoms. It is another object of the present invention to use a rich data set over multiple nights to titrate the CPAP treatment. It is another object of the present invention to use a closed-loop or partially closed-loop system to automatically titrate the CPAP treatment based on the subject's physiological signals. It is still another object of the present invention to treat a subject's apnea in a predictive manner. It is still another object of the present invention to provide a system or method of treating a subject's apnea using the subject's physiological signals. It is still another object of the present invention to provide a device and method of titration in the subject's home. It is still another object of the present invention to provide a device and method of titration in the hospital's acute or sub-acute settings, such as for postoperative management of care.


1. Field of the Invention
This invention relates to electronic circuits and, more particularly, to clock generation and timing circuits.
2. Description of the Related Art
The following descriptions and examples are given as background only.
Phase locked loops (PLLs) are commonly used for data and telecommunications, frequency synthesis, clock recovery, and similar applications. In some cases, a PLL may be used in the I/O interfaces of digital integrated circuits to hide clock distribution delays and to improve overall system timing. In other cases, a PLL may be used as a clock multiplier for downstream circuit components. In one example application, an input clock of 10 Mhz can be multiplied by the PLL to yield a 1000 Mhz output signal, which may then be used for clocking internal circuit components. Ideally, this input (or source) clock multiplication could result in an output that is in perfect phase alignment with the input clock. However, in practice this is often not the case.
A typical PLL device 100 is shown in FIG. 1 as including an optional reference divider (div M) 120, a phase frequency detector (PFD) 130, a charge pump 140, a low pass filter 150, a voltage controlled oscillator (VCO) 160, and an optional frequency divider (div N) 170.
During operation, PLL circuit 100 receives a reference clock signal (FREF) from an external source (e.g., a crystal oscillator) 110. The phase frequency detector compares the reference signal (FREF) to a feedback signal (FVCO) generated by components within the PLL circuit. More specifically, PFD 130 detects differences in frequency and/or phase between the reference and feedback clock signals, and generates compensating “up” and “down” signals in response thereto. The particular control signals generated depend on whether the feedback clock signal is lagging or leading the reference clock signal in frequency or phase. The up/down control signals are passed through charge pump 140 and filter 150 to integrate the control signals into a control voltage, which is sent to the VCO. The voltage-controlled oscillator converts the voltage information into one or more output frequencies (FVCO). One of these output frequencies may then be sent back to the PFD via a feedback loop.
In some cases, frequency divider 170 and reference divider 120 may be included for adjusting the frequencies of the feedback and reference clock signals, respectively. For example, frequency divider 170 may be used for dividing the frequency of a VCO output signal (FOUT) to produce a divided down feedback signal (FOUT/N), while reference divider 120 is used for dividing the frequency of the external clocking signal to produce a divided down reference signal (FREF/M), which is similar or dissimilar to the divided down feedback signal. In such cases, PLL 100 may function as a clock or frequency multiplier. However, dividers 120 and/or 170 may not be included in all cases.
In conventional PLL devices, high-performance inductor-capacitor (LC) VCOs are often used to create low-noise, high-speed PLLs. For example, LC-type oscillators are often used in PLL designs tailored for wireless and low power applications, as well as other applications requiring precise timing. Unfortunately, LC-type oscillators have a tight frequency range, which is sensitive to variations in process, voltage and temperature (PVT). In some cases, variations in PVT may cause the VCO frequency range to shift outside of a target range, thereby limiting the usable frequency range of the PLL device. For this reason, various solutions have been proposed for extending the usable frequency range of a PLL device employing an LC-type oscillator.
One solution to the above-mentioned problem is to create or use a VCO with a wide frequency range. For example, a wider frequency range can be obtained by increasing the gain of an LC-type oscillator, or by using a completely different oscillator (e.g., a ring oscillator) with an inherently wider frequency range. Unfortunately, large VCO gains are undesirable because of their sensitivity to noise. In addition, although a wide frequency range may be easily obtained when the ring oscillator is employed as a VCO, the ring oscillator is not without limitations and usually demonstrates poorer phase-noise performance than the LC-type oscillator.
Another solution is to use multiple LC-type oscillators within the PLL device. For example, a first VCO may be used for generating frequencies within a 2-2.45 GHz range, while a second VCO is used for generating frequencies within a 2.45-2.5 Ghz range. Additional or alternative VCOs may be used for generating frequencies within other ranges. Unfortunately, the second solution may be undesirable in many applications, due to the relatively large die area consumed by the additional VCO(s).
In yet another solution, an LC-type VCO may be calibrated during the manufacturing test process to shift the VCO operating frequency into a desired range. For example, test circuitry may be used for measuring the max VCO frequency, measuring the min VCO frequency and calculating an average of the two. The test circuitry may then be used for adjusting the programmable bits supplied the LC-type VCO until the measured value(s) equal a set of target value(s). Unfortunately, the third conventional solution is too slow (i.e., adds additional test time) and does not compensate for the environmental conditions that the device will actually be used in (i.e., the method does not account for power supply and temperature variations, only for process variations).
In yet another solution, an LC-type VCO may be calibrated during operation of the PLL device to shift the VCO operating frequency into a desired range. One such solution is described in a paper entitled “A Delta-Sigma Fractional-N Synthesizer using a Wide-Band Integrated VCO and a Fast AFC Technique for GSM/GPRS/WCDMA Applications,” and published in the July 2004 issue of IEEE Journal of Solid-State Circuits (JSSCC), vol. 39, no. 7, pgs. 1164-1169. In this solution, a pair of switches are used for disconnecting the low pass filter from the VCO (i.e., opening the loop) to enter a VCO calibration mode. To achieve calibration, digital frequency counters are used for counting the number of reference and feedback clock pulses supplied thereto. A comparator is then used to determine the proper VCO frequency by comparing the number of feedback pulses to the number of reference pulses, while a state machine is used for adjusting the programmable bits supplied to the VCO. Unfortunately, the frequency counters and state machine used in the fourth solution are too slow for many high-speed applications (e.g., it may take about 650 clock cycles to complete the calibration using the method describe above). In addition, the switches used for disconnecting the low pass filter from the VCO introduce a series resistance into the PLL signal path. This is undesirable because it tends to alter the behavior of the filter.
Therefore, a need remains for an improved calibration solution that may be used for extending the usable frequency range of a PLL device. Preferably, the improved solution would allow frequency calibration and compensation of a narrow band PLL over a wide range of actual environmental conditions, including process, voltage, and temperature. Even more preferably, the improved solution would provide a circuit and method for calibrating an LC-type oscillator during operation of a PLL device, wherein said calibration is performed without opening the loop and with much greater speed than possible with conventional solutions.


I. Field
The present invention relates generally to electronics circuits, and more specifically to current sources and active circuits.
II. Background
Current sources are widely used to provide current for various circuits such as amplifiers, buffers, oscillators, and so on. Current sources may be used as bias circuits to provide bias currents, active loads to provide output currents, and so on. Current sources are often fabricated on integrated circuits (ICs) but may also be implemented with discrete circuit components.
As IC fabrication technology continues to improve, the size of transistors continues to shrink. The smaller transistor size enables more transistors and thus more complicated circuits to be fabricated on an IC die or, alternatively, a smaller die to be used for a given circuit. The smaller transistor size also supports faster operating speed and provides other benefits.
Complementary metal oxide semiconductor (CMOS) technology is widely used for digital circuits and many analog circuits. A major issue with shrinking transistor size in CMOS is leakage current, which is the current passing through a transistor when it is turned off. A smaller transistor geometry results in higher electric field (E-field), which stresses a transistor and causes oxide breakdown. To decrease the E-field, a lower power supply voltage is often used for smaller geometry transistors. However, the lower supply voltage also increases the propagation delay of the transistors, which is undesirable for high-speed circuits. To reduce the delay and improve operating speed, the threshold voltage (Vt) of the transistors is reduced. The threshold voltage determines the voltage at which the transistors turn on. However, the lower threshold voltage and smaller transistor geometry result in higher leakage current.
Leakage current is more problematic as CMOS technology scales smaller. This is because leakage current increases at a high rate with respect to the decrease in transistor size. Leakage current can impact the performance of certain circuits such as phase lock loops (PLLs), oscillators, digital-to-analog converters (DACs), and so on.
Some common techniques for combating leakage current include using high threshold voltage (high-Vt) transistors and/or larger transistor sizes (e.g., longer gate lengths). High-Vt transistors may impact circuit performance (e.g., slower speed) and typically require an additional mask step in the IC fabrication process. Larger-size transistors are marginally effective at combating leakage current since (1) leakage current is a relatively weak function of channel length and (2) there are practical limits on how long the channel length may be extended. Both of these solutions may thus be inadequate for certain circuits.
There is therefore a need in the art for a current source with low leakage current and good performance.


The semiconductor industry has continually sought ways to produce memory devices with an increased number of memory cells per memory die. In non-volatile memory (e.g., NAND flash memory), one way to increase memory density is by using a vertical memory array, which is also referred to as a three-dimensional (3-D) memory array. One type of vertical memory array includes semiconductor pillars that extend through openings (e.g., holes) in layers of conductive material (also referred to as word line plates or control gate plates), with dielectric materials at each junction of the semiconductor pillars and the conductive materials. Thus, multiple transistors can be formed along each pillar. Vertical memory array structures enable a greater number of transistors to be located in a unit of die area by building the array upwards (e.g., vertically) on a die, as compared to structures with traditional planar (e.g., two-dimensional) arrangements of transistors.
Vertical memory arrays and methods of forming them are described in, for example: U.S. Patent Application Publication No. 2007/0252201 of Kito et al., now U.S. Pat. No. 7,936,004, issued May 3, 2011; Tanaka et al., “Bit Cost Scalable Technology with Punch and Plug Process for Ultra High Density Flash Memory,” Symposium on VLSI Technology Digest of Technical Papers, pp. 14-15 (2007); Fukuzumi et al., “Optimal Integration and Characteristics of Vertical Array Devices for Ultra-High Density, Bit-Cost Scalable Flash Memory,” IEDM Technical Digest, pp. 449-52 (2007); and Endoh et al., “Novel Ultrahigh-Density Flash Memory with a Stacked-Surrounding Gate Transistor (S-SGT) Structured Cell,” IEEE Transactions on Electron Devices, vol. 50, no. 4, pp. 945-951 (April, 2003).
Conventional vertical memory arrays require an electrical connection between the conductive materials (e.g., word line plates or control gates) and access lines (e.g., word lines) so that memory cells in the 3-D array may be uniquely selected for writing or reading functions. One method of forming an electrical connection includes forming a so-called “stair-step” structure at the edge of the conductive materials. FIGS. 1A through 1D show one conventional method of creating a stair-step structure 10 in a stack of conductive materials 12. As shown in FIG. 1A, conductive materials 12 are separated by insulating materials 14 between the conductive materials 12. A mask 16 (e.g., photoresist material) is formed over the topmost insulating material 14 and patterned to expose a portion of the insulating material 14a, the exposed portion having a width of one so-called “step” of the stair-step structure 10 to be formed. An anisotropic etch 18, such as a reactive ion etch (RIE) or other dry etch, is performed to remove the insulating material 14a at the portion exposed through the mask 16. The pattern in the insulating material 14a is then transferred to the conductive material 12a. The exposed insulating material 14a is removed by one dry etch process that stops on the conductive material 12a, and the exposed conductive material 12a is then removed by another dry etch process that stops on the insulating material 14b. Next, the mask 16 is reduced in size by removing a portion of the mask (also known as “trimming”), such as by isotropic etching, to expose another portion of the insulating material 14a, as shown in FIG. 1B.
The process is repeated by subjecting the structure to an anisotropic etch 18, including removing exposed portions of the two insulating materials 14a and 14b and subsequently removing exposed portions of the two conductive materials 12a and 12b. As shown in FIG. 1C, the successive reduction in size of the mask 16 and the repeated dry etch processes are continued until the insulating material 14c and conductive material 12c is exposed, the mask 16 is removed, and a stair-step structure 10 remains. Word line contacts 20 are formed to extend through each respective insulating material 14 and electrically contact each conductive material 12, as shown in FIG. 1D. The top of each word line contact 20, as viewed in FIG. 1D, connects to a conductive word line (not shown). While FIGS. 1A through 1D illustrate using two anisotropic etches 18 to create three so-called “steps” of the stair-step structure 10, the acts of etching the insulating material 14, etching the conductive material 12, and trimming the mask 16 may be repeated to create more steps (and thus contact regions for word line contacts). Current conventional methods have been used to form more than eight contact regions (e.g., steps).
As the desired number of steps in the conventional stair-step structure increases, the margin of error associated with each act in the process of forming the steps correspondingly decreases when using the conventional method. For example, and as explained above, each iteration of the conventional method includes trimming the mask, etching the insulating material, and etching the conductive material. The desired number of steps is formed by repeating these acts as many times as the number of conductive materials in the stack. Each act of the conventional method has an associated etch control error because the size of each step is designed to fall within a particular range (e.g., tolerance) to allow enough room for a contact to be formed thereon while keeping the overall size of the stair-step structure small. Additionally, the relative locations of the steps are designed to fall within a range of locations in order to accurately form contacts thereon. As the number of iterations increases, any deviation from a target step width or location may be compounded because errors in one material are transferred to an underlying material. For a high number of steps in the stair-step structure, the margin of error to be achieved for the etch rate control may be less than one percent (1%). Small margins of error are difficult and costly to attain using conventional methods. Furthermore, because the mask is repeatedly trimmed, the method may start with a mask of high thickness, which may be difficult to repeatedly pattern and trim with the precision needed to have the necessary control over step width. Furthermore, the large amount of mask material is expensive and time-consuming to both form and remove.
Space savings in a memory device incorporating a vertical memory array may be accomplished by reducing the area that a stair-step structure covers. One method of reducing this area is described in U.S. Patent Application Publication No. 2009/0310415 to Jin et al., now U.S. Pat. No. 8,325,527, issued Dec. 4, 2012. Although some space is saved by aligning the word line contacts in the same direction as the bit lines, further improvements and reductions in cost in the manufacturing of such structures, as well as alternative methods of reducing the area covered by the stair-step structures, are desired. For example, the method described in Jin et al. uses a unique mask for each etch act to form the steps, which adds significant cost because of a high number of photolithographic reticles used to form the masks. Reductions in cost and improvements in controllability of manufacturing stair-step structures are, therefore, desired.


1. Field of the Invention
The present invention is related to dynamic random access memory (DRAM) devices. In particular, it relates to a system and method for improving a DRAM""s ability to capture data correctly on all data paths (DQs).
2. Description of Related Art
When double data rate (DDR) is applied to a memory device, the data is input/output to/from the memory device on both edges (i.e., rising and falling) of an external system clock CLK. That is, the memory device (e.g., a DRAM) receives/outputs two bits of data per whole CLK cycle, called 2-bit (or 2n) prefetch. A DDR II memory device is similar to a DDR device but runs off an external clock CLK which is twice as fast (e.g., 200 MHz) as a DDR external clock. DDR II uses a 4-bit (or 4n) prefetch such that 4-bits are transferred in/out of a data path then handled as one 4-bit-wide piece of data inside the memory.
Double data rate memory is one key element for boosting memory device throughput to keep pace with the ever-increasing throughput performance of microprocessors. By doubling the memory bandwidth over current generation synchronous DRAM devices, DDR II can provide a cost-effective, high-performance main memory solution that does not require a significant development or manufacturing investment, while maintaining a cost structure consistent with synchronous DRAMs. DDR II memory devices and modules are well-suited for a broad range of applications, especially the workstation and server markets, where the high module density and device architecture can meet the performance and reliability demands of these products.
DDR II memory technology has been defined and standardized by the Joint Electron Devices Engineering Council (JEDEC) as a next-generation memory solution. This technology is intended to facilitate adoption in a wide range of products, and to be offered in both device and module form from all major suppliers.
The DDR II standard proposes to have a minimum respective data setup and hold times of approximately 0.25 ns. Referring to FIG. 1, the data setup time is defined as the time the data is at the data input/output pin (DQ) before the next external system clock CLK edge arrives. The hold time is defined as the time the data stays valid after the clock edge. The entire window of data (setup and hold) is referred to as the xe2x80x9cdata eye.xe2x80x9d A data eye of approximately 1.5 ns is required in order for the DRAM to reliably capture the data (i.e., the DQ input pulse width (DIPW)). Moreover, in order for the DRAM to reliably capture the data, the clock CLK edge (either rising or falling) must be centered on the data eye. If the clock edge is not centered on the data eye, either one of the setup or hold times will be in danger of being of an insuffliciently short duration, which may prevent the DRAM from properly capturing the data.
There are many reasons why the clock edge may not be centered on the data eye. Some of those reasons include clock jitter, noise on the board, different lengths of data traces on the board, etc., which cause a clock skew to occur. The problem is exacerbated with higher clock frequencies since the setup and hold time is reduced in those cases, thus, leaving less room for error when attempting to locate the clock edge on the center of the data eye.
One overly complex solution to maintaining the clock edge near the center of the data eye that has been proposed is known as xe2x80x9cSyncLink.xe2x80x9d The SyncLink approach actually builds a mathematical model of the entire data eye and determines the exact location of the leading and trailing edges of the data. Once the system has calculated the location of the leading and trailing edges, the center of the data eye is calculated and the clock edge is located at the center of the data eye. While this method has proven to be accurate, it has also proven to be unnecessarily complex for most DDR II operating environments. This complexity results in greater chip complexity and cost. Thus, a simplified system and method are required to ensure that data which is input to a DRAM is properly timed relative to the external DRAM clock CLK such that the data is driven by a clock edge located on the center of the data eye and is, thereby, properly captured by the DRAM.
The present invention provides a simple system and method for controlling the rate with which data is input to a DRAM relative to the external DRAM clock CLK such that its clock edges are always centered on the data eye of input data to the DRAM, thus, ensuring that the data may always be properly captured by the DRAM.
In accordance with an exemplary embodiment of the invention, upon power-up or reset, during the existing initialization cycle time of the DRAM (i.e., 200 cycles under the DDR II standard), the write data capture calibration method of the invention is performed. A write command is issued by a memory controller and a simple repetitive string of data such as e.g., xe2x80x9c110011001100 . . . xe2x80x9d is sent by the memory controller to all the write data input paths (DQs) of the memory device, along with a bi-directional strobe (DQS) that is essentially a data clock aligned with data when it leaves the controller. The DRAM captures the data and performs a DQ to DQS delay adjustment to center the clock edge on the data eye for each DQ path. In the meantime, a delay lock loop (DLL) compares an internal data clock with the external clock to ensure they are both in phase. If they are out of phase, the DLL delays the internal clock until the two clocks are in phase. An auto-refresh automatically cancels the write data capture calibration method upon completion of the initial 200 cycles and the DRAM is returned to normal operation.


To meet the demand for wireless data traffic, which has increased since deployment of 4th-generation (4G) communication systems, efforts have been made to develop an improved 5th-generation (5G) or pre-5G communication system. Therefore, the 5G or pre-5G communication system is also called a ‘beyond 4G network’ or a ‘post long-term evolution (LTE) system’.
It is considered that the 5G communication system will be implemented in millimeter wave (mmWave) bands, e.g., 60 GHz bands, so as to accomplish higher data rates. To reduce propagation loss of radio waves and increase a transmission distance, a beam forming technique, a massive multiple-input multiple-output (MIMO) technique, a full dimensional MIMO (FD-MIMO) technique, an array antenna technique, an analog beam forming technique, and a large scale antenna technique are discussed in 5G communication systems.
In addition, in 5G communication systems, development for system network improvement is under way based on advanced small cells, cloud radio access networks (RANs), ultra-dense networks, a device-to-device (D2D) communication, a wireless backhaul, a moving network, a cooperative communication, coordinated multi-points (CoMP), reception-end interference cancellation, and the like.
In the 5G system, a hybrid frequency shift keying (FSK) and Feher's quadrature amplitude modulation (FQAM) and a sliding window superposition coding (SWSC) as an advanced coding modulation (ACM) scheme, and a filter bank multi carrier (FBMC) scheme, a non-orthogonal multiple access (NOMA) scheme, and a sparse code multiple access (SCMA) scheme as an advanced access technology have been developed.
A communication system supporting a full-duplex scheme may increase system capacity doubles by performing a transmitting operation and a receiving operation on the same frequency at the same time.
In the communication system supporting the full-duplex scheme, there is a self-interference (SI) signal which occurs since a signal transmitted by a transmitting device is received in the transmitting device due to a characteristic of the full-duplex scheme.
So, various schemes for canceling the SI signal have been proposed in the communication system supporting the full-duplex scheme, and a typical one is a scheme for cancelling an SI signal which affects a receiving circuit of a transmitting device in a circuit domain.
The scheme for cancelling the SI signal in the circuit domain may be classified into a digital SI cancellation (SIC) scheme and an analog SIC scheme according to whether magnitude of the SI signal is within a digital dynamic range that the SI signal may be received in a digital domain. The digital SIC scheme denotes a scheme for cancelling an SI signal using a digital signal processing scheme, and the analog SIC scheme denotes a scheme for cancelling an SI signal using both an analog circuit and a digital signal processing scheme. It is general to use the digital SIC scheme and the analog SIC scheme at the same time for cancelling an SI signal in a system level.
Each of the digital SIC scheme and the analog SIC scheme will be described below.
Firstly, the analog SIC scheme will be described below.
In the analog SIC scheme, it will be assumed that a received SI signal includes a finite number of signals received after fixed delay time which a transmitting device already knows after the transmitting device transmits a signal. Under this assumption, the analog SIC scheme may adjust a gain for a transmission signal which is divided from an analog transmitting circuit included in the transmitting device, and cancel an SI signal from a signal received from the transmitting device by applying a circuit with fixed delay time. Here, the gain adjusted through the circuit may be acquired based on an interference characteristic estimated by the transmitting device.
Secondly, the digital SIC scheme will be described below.
The digital SIC scheme detects a channel characteristic from difference between a transmission signal and a reception signal using a signal divided from a digital transmission signal of a transmitting device and a digital reception signal which is received using a digital signal processing scheme in the transmitting device, and cancels an SI signal from the digital reception signal by applying the channel characteristic in reverse.
The digital SIC scheme may cancel only an SI signal which is within a digital dynamic range. So, an SI signal which is not within the digital dynamic range may be cancelled only after an analog SIC operation is performed.
So, a performance of a communication system supporting a full-duplex scheme which uses the digital SIC scheme is determined according to a performance of the analog SIC scheme.
The analog SIC scheme is implemented with a scheme for previously predicting the number of SI signals which are received in a transmitting device after being reflected from the transmitting device, and adding an analog circuit which may cancel each SI signal.
The analog SIC scheme may not cancel an SI signal if a characteristic of the SI signal is different from a characteristic of an SI signal predicted by the analog circuit, or the number of SI signals is greater than the number of SI signals predicted by the analog circuit due to various reasons such as a situation that an SI signal is received from an outside of a transmitting device, and the like.
The above information is presented as background information only to assist with an understanding of the present disclosure. No determination has been made, and no assertion is made, as to whether any of the above might be applicable as prior art with regard to the present disclosure.