Market adoption of wireless LAN (WLAN) technology has exploded, as users from a wide range of backgrounds and vertical industries have brought this technology into their homes, offices, and increasingly into the public air space. This inflection point has highlighted not only the limitations of earlier-generation systems, but the changing role WLAN technology now plays in people's work and lifestyles, across the globe. Indeed, WLANs are rapidly changing from convenience networks to business-critical networks. Increasingly users are depending on WLANs to improve the timeliness and productivity of their communications and applications, and in doing so, require greater visibility, security, management, and performance from their network.
The rapid proliferation of lightweight, portable computing devices and high-speed WLANs enables users to remain connected to various network resources, while roaming throughout a building or other physical location. The mobility afforded by WLANs has generated a lot of interest in applications and services that are a function of a mobile user's physical location. Examples of such applications include: printing a document on the nearest printer, locating a mobile user or rogue access point, displaying a map of the immediate surroundings, and guiding a user inside a building. The required or desired granularity of location information varies from one application to another. Indeed, the accuracy required by an application that selects the nearest network printer, or locates a rogue access point, often requires the ability to determine in what room a wireless node is located. Accordingly, much effort has been dedicated to improving the accuracy of wireless node location mechanisms.
The use of radio signals to estimate the location of a wireless device or node is known. For example, a Global Positioning System (GPS) receiver obtains location information by triangulating its position relative to four satellites that transmit radio signals. The GPS receiver estimates the distance between each satellite based on the time it takes for the radio signals to travel from the satellite to the receiver. Signal propagation time is assessed by determining the time shift required to synchronize the pseudo-random signal transmitted by the satellite and the signal received at the GPS receiver. Although triangulation only requires distance measurements from three points, an additional distance measurement from a fourth satellite is used for error correction.
The distance between a wireless transmitter and a receiver can also be estimated based on the strength of the received signal, or more accurately the observed attenuation of the radio signal. Signal attenuation refers to the weakening of a signal over its path of travel due to various factors like terrain, obstructions and environmental conditions. Generally speaking, the magnitude or power of a radio signal weakens as it travels from its source. The attenuation undergone by an electromagnetic wave in transit between a transmitter and a receiver is referred to as path loss. Path loss may be due to many effects such as free-space loss, refraction, reflection, and absorption.
In business enterprise environments, most location-tracking systems are based on RF triangulation or RF fingerprinting techniques. RF triangulation calculates a mobile user's location based upon the detected signal strength of nearby access points (APs). It naturally assumes that signal strength is a factor of proximity, which is true a majority of the time. However, the multipath phenomenon encountered in indoor RF environments does present certain difficulties in locating wireless nodes, since reflection and absorption of RF signals affects the correlation between signal strength and proximity. RF fingerprinting compares a mobile station's view of the network infrastructure (i.e., the strength of signals transmitted by infrastructure access points) with a database that contains an RF physical model of the coverage area. This database is typically populated by either an extensive site survey or an RF prediction model of the coverage area. For example, Bahl et al., “A Software System for Locating Mobile Users: Design, Evaluation, and Lessons,” http://research.microsoft.com/˜bahl/Papers/Pdf/radar.pdf, describes an RF location system (the RADAR system) in a WLAN environment, that allows a mobile station to track its own location relative to access points in a WLAN environment.
The RADAR system relies on a so-called Radio Map, which is a database of locations in a building and the signal strength of the beacon packets emanating from the access points as observed, or estimated, at those locations. For example, an entry in the Radio Map may look like (x, y, z, ssi (i=1 . . . n)), where (x, y, z) are the physical coordinates of the location where the signal is recorded, and ssi is the signal strength of the beacon signal emanating from the ith access point. According to Bahl et al., Radio Maps may be empirically created based on heuristic evaluations of the signals transmitted by the infrastructure radios at various locations, or mathematically created using a mathematical model of indoor RF signal propagation. To locate the position of the mobile user in real-time, the mobile station measures the signal strength of each of access points within range. It then searches a Radio Map database against the detected signal strengths to find the location with the best match. Bahl et al. also describe averaging the detected signal strength samples, and using a tracking history-based algorithm, to improve the accuracy of the location estimate. Bahl et al. also address fluctuations in RF signal propagation by using multiple Radio Maps and choosing the Radio Map which best reflects the current RF environment. Specifically, one access point detects beacon packets from other access points and consults a radio map to estimate its location, and evaluates the estimated location with the known location. The RADAR system chooses the Radio Map which best characterizes the current RF environment, based on a sliding window average of received signal strengths.
While the RADAR system works for its intended objective, even in this system, location accuracy decreases with the error in detecting the strength of RF signals. For example, individual differences as to how two different wireless nodes detect and report signal strength can cause errors in location, since the Radio Maps assume no error in such measurements. Accordingly, two wireless nodes in the same location that detect different signal strengths will compute different estimated locations. Still further, while the RADAR system allows a mobile station to track its own location, it does not disclose a system that allows the WLAN infrastructure to track the location of wireless nodes, such as rogue access points. Such a system is desirable as it obviates the need for special client software to be installed on the mobile stations.
This paradigm shift, however, presents certain problems. As discussed above, the Radio Maps in the RADAR system are constructed from the point of view of a wireless node in an RF environment that includes access points in known locations. In other words, the Radio Maps are constructed based on heuristic and/or mathematical evaluations of the propagation of signals from the access points to a wireless node at a given location. Accordingly, the RADAR system need not assume symmetry of path loss between a given location and the access points in the RADAR system, since the mobile station detects the signal strength of the access points and computes its own location. In addition, since the location of a wireless node is based on path loss, the transmit power of the radio transmitters used to determine location must also be known. In the RADAR system, this is not problematic, since the signals used to determine location are transmitted by access points, whose transmit power can be controlled or easily determined. Estimating location based on signals transmitted by a wireless node, however, can be problematic, since transmit power can vary among wireless device manufacturers, and/or may be individually configured by the mobile user. Furthermore, the more complicated issue of variable transmit power exists not only due to the device or intended application itself, but also due to placement of the wireless node (e.g., a RFID tag, etc.) placed among or inside of other objects, or located within or behind various physical barriers or obstructions.
One approach to this problem is to assume symmetry in path loss between a given location in an RF environment and the radio transceivers used to detect signals transmitted by the wireless nodes. Furthermore, these approaches also assume a uniform transmit power for the wireless nodes in light of the fact that legal regulations, as well as current chip set technology, generally places an upper limit on transmit power. These two assumptions, however, can significantly impact the accuracy of locating a wireless node. As discussed above, the RADAR system, for example, finds the location coordinates in the Radio Map that are the best fit based on the detected signal strengths. That is, for each point in the Radio Map, the location metric computes the Euclidean distance between the detected signal strength values and the values in the Radio Map.
The following equation provides an illustrative example, assume for didactic purposes that a given wireless node is detected by three access points. The signal strength samples are RSSIap1 RSSIap2, and RSSIap3, while the RF coverage maps for each of the access points are denoted as MAPap1, MAPap2, MAPap3, where the coverage maps include access point signal strength values detected or computed for different locations in a defined region. Again, assuming path loss symmetry and a uniform transmit power, individual error surfaces for each access point can be created based on the signal strength detected at each access point, (RSSIap1, etc.) and the signal strength values in the individual coverage maps (e.g., MAPap1, etc.). That is, the error surface is the difference between the observed signal strength at a given access point less the signal strength values in the coverage map. The locations in this coverage map where the difference is zero are the most likely locations. In many situations, however, the measured signal strengths, RSSIap1 RSSIap2, and RSSIap3, do not match the signal strengths recorded in the coverage maps MAPap1, MAPap2, MAPap3 at any one location. In this case, it is desirable to find the location that is “closest” to matching RSSIap1 RSSIap2, and RSSIap3—in other words, the location that minimizes some function of MAPap1, MAPap2, MAPap3, RSSIap1 RSSIap2, and RSSIap3. Bahl et al., supra, describe several ways in which this function is created, including minimum mean squared error, minimum distance, and minimum Manhattan grid distance. Furthermore, a total error surface, ErrSurf, can be computed based on the sum of the squares (to neutralize positive and negative differences) of the individual error surfaces (i.e., the difference between the detected signal strength values and the signal strength values in each coverage maps), as follows:ErrSurf=[(RSSIap1−MAPap1)^2+(RSSIap2−MAPap2)^2+(RSSIap3−MAPap3)^2]/3In one implementation, the estimated wireless node location is derived from the minimum or minimum of this total error surface.
However, a change in the wireless node's effective transmit power (or, in the RADAR system, inaccuracies in detecting signal strength by the wireless nodes) will adversely affect the accuracy of this metric. For example, an N dB difference between the actual and assumed transmit power of a wireless node would cause a N dB change in the detected signal strengths. Rather than merely shifting the individual signal strength differences for each point in the individual error surfaces up by some fixed amount, the individual differences between the detected signal strengths and the signal strength values in the error surface can change quite dramatically. Indeed, each point in the individual error surfaces are shifted an amount proportional to the dB error. This circumstance moves some areas of the total error surface up relative to others, and some areas of the total error surface down relative to others, significantly altering the shape of the error surface, as well as the location, shape, and size of its minima. It also creates unpredictable error with changes in transmit power. Similar problems will occur for a fixed error in the “link or path loss symmetry” where the path loss from access point to wireless node differs from the path loss from wireless node to access point by some fixed amount due to propagation characteristics, vantage point, and the like. In addition, sources of RF interference typically have unknown transmit powers, and may only partially overlap the frequency band in which wireless nodes operate. Estimating the location of these interference sources requires a method that does not depend entirely on the absolute detected signal strength value.
In light of the foregoing, a need in the art exists for a wireless node location mechanism that reduces the errors in computing the location of a wireless node due to commonly occurring circumstances, such as variations in wireless node transmit power, errors in signal strength detection, and/or direction-dependent path loss. Embodiments of the present invention substantially fulfill this need.


In a trade show, an exhibition or a similar environment, in order to help participants to get the most out of the visit, there are usually some electronic tour-guide (purchase guide) systems provided which allow users to browse or hear more information of an object they are interest in through a portable client device. Based on the way these systems decide of which object the user is interest in, generally they can be divided into two types. In the first type of systems, the user should manually key in the identification (ID) of the object he/she is interest in on his/her client device. However, this is quite troublesome for users. The second type of systems is the so-called location-based system, where some kind of location detection mechanism (such as radio frequency identification (RFID)) is used, and the client device will automatically deliver the information of an object to the user when the user is standing approximate to this object. Although this type of system is more convenient than the first one, it has following drawbacks. First, when there are several objects with same or similar distance and directional angle to the user, there is no reliable way to decide which object should be the one the user is interested in. Second, after all, the location approximation has no certain relation with a user's real interest. For example, when the user is facing and looking at a picture hanged on the wall several steps away while there is another exhibited item just next to the user's side, in this case, the object closest to the user is not the one the user is actually interested in. So, the object information delivered to the user by a location-based system is not the information of the object the user is actually interested in.


The inventive concept relates to integrated circuit(s), and more particularly, to computer-implemented method(s) for designing and/or manufacturing integrated circuit(s).
The design and manufacture of integrated circuits is a highly complex process. The evolution of integrated circuits is characterized by an increasing integration density of the constituent elements. In order to obtain a reliable integrated circuit at the end of the design and manufacturing process, a number of design constraints (e.g., element proximity, signal timing, power consumption, etc.) are usually defined in order to protect the integrity of the overall design. As integration density increases, the possibility of violating one or more design constraint(s) rises.
The evolution of integrated circuits is also characterized by a desire to reduce or minimize the physical size of and/or the power consumed by the integrated circuit (or the incorporating semiconductor chip). Unfortunately, as the number of design constraints for an integrated circuit increases, corresponding efforts to satisfy the design constraints tend to drive up physical size and/or power consumption. For example, efforts to satisfy a particular timing constraint for an integrated circuit may result in the addition of a buffer cell (or analogous delay element) in order to improve hold time margin(s). Alternately or additionally, efforts to satisfy a particular noise constraint for an integrated circuit may result in the addition of a decoupling capacitor. Such efforts tend to increase the physical size and/or power consumption of the integrated circuit.


The present invention relates to compositions and methods for the prevention and treatment of neoplastic and oncogenic disorders, with combinations of certain limonoids, flavonoids and/or tocotrienols. Limonoids are a group of chemically related triterpene derivatives found in the Rutaceae and Meliaceae families. Limonoids are among the bitter principals in citrus juices such as lemon, lime, orange and grapefruit. Flavonoids are polyphenolic compounds that occur unbiquitously in plant foods especially in orange, grapefruit and tangerine. Tocotrienols are present in palm oil and are a form of vitamin E having an unsaturated side chain. In the practice of the cancer prevention and/or treatment of the invention the limonoids, flavonoids and tocotrienols are used to inhibit the development progression and proliferation of cancer cells. Preferred compositions of the invention are those which specifically or preferentially prevent transformation of preneoplastic cells to tumor cells, and prevent or inhibit tumor cell proliferation, invasion and metastasis without general cytotoxic effects.
2.1 Limonoids
Limonoids are a group of chemically related triterpene derivatives found in the Rutaceae and Meliaceae families. Limonoids are among the bitter principles found in citrus fruits such as lemons, lime, orange and grapefruit. They are also present as glucose derivatives in mature fruit tissues and seed, and are one of the major secondary metabolites present in Citrus. Limonoids have been found to have anti-carcinogenic activity in laboratory animals. The furan moiety attach to the D-ring is specifically responsible for detoxifying of the chemical carcinogen glutathione-S-transferase enzyme system (Lam, et al., 1994, Food Technology 48:104-108).
Citrus fruit tissues and byproducts of juice processing such as peels and molasses are sources of limonoid glucosides and citrus seed contain high concentrations of both limonoid aglycones and glucosides. Limonoid aglycones m the fruit tissues gradually disappear during the late stages of fruit growth and maturation.
Thirty-eight limonoid aglycones have been isolated from Citrus. The limonoids are present in three different forms: the dilactone (I) is present as the open D-ring form (monolactone), the limonoate A-ring lactone (II) and the glucoside form (III). Only the monolactones and glucosides are present in fruit tissues. (Hasegawa S. et al., 1994, in Food Phytochemicals for Cancer Prevention I, eds M-T. Huang et al, American Chemical Society, 198-207). ##STR1##
Compound III is the predominant limonoid glucoside found in all juice samples. In orange juice it comprises 56% of the total limonoid glucosides present, while in grapefruit and lemon juices, it comprises an average of 63% to 66% respectively. Procedures for the extraction and isolation of both aglycones and glucosides have been established to obtain concentrated sources of various limonoids (Lam, L. K. T. et al., 1994, in Food Phytochemicals for Cancer Prevention, eds. M. Huang, T. Osawa, C. Ho and R. T. Rosen, ACS Symposium Series 546, p 209). The use of limonoids in combination with a flavonoid, tocotrienol, a cancer chemotherapeutic agent, or a combination of any one of these agents, has not been reported for the prevention and treatment of neoplastic diseases.
2.2 Flavonoids.
Epidemiological studies have shown that flavonoids present in the Mediterranean diet may reduce the risk of death from coronary heart disease (Hertog, M. G. et al., 1993, Lancet: 342, 1007-1011). Soybean isoflavones for example, genistein, which is a minor component of soy protein preparations may have cholesterol-lowering effects (Kurowska, E. M. et al., 1990, J. Nutr. 120:831-836). The flavonoids present in citrus juices such as orange and grapefruit include, but are not limited to, hesperetin and naringenin respectively.
##STR2## 5 7 3' 4' HESPERETIN OH OH OH OCH.sub.3 NARINGENIN OH OH -- OH
The flavonoids preset in tangerine include, but are not limited to tangerevtin or nobiletin. These flavonoids were found to inhibit growth of both estrogen receptor-negative (ER-) and positive (ER+) breast cancer cells in culture and act synergistically with tamoxifen and tocotrienols (Guthrie N. et aL, 1996, Proc. Am. Inst. Cancer Res., Abs.#8).
 ##STR3## 5 6 7 8 4' 5' TANGERETIN O CH.sub.3 O CH.sub.3 O CH.sub.3 O CH.sub.3 O CH.sub.3 -- NOBILETIN O CH.sub.3 O CH.sub.3 O CH.sub.3 O CH.sub.3 O CH.sub.3 O CH.sub.3
2.3 Tocotrienols in Palm Oil
Tocotrienols are present in palm oil and are a form of vitamin E having an unsaturated side chain. They include, but are not limited to alpha-tocotrienol, gamma-tocotrienol or delta-tocotrienol.
 ##STR4## R1 R2 R3 .alpha.-tocotrienol CH.sub.3 CH.sub.3 CH.sub.3 .gamma.-tocotrienol H CH.sub.3 CH.sub.3 .delta.-tocotrienol H H CH.sub.3
2.4 Cancer Growth and Chemotherapy
Cancer is a disease of inappropriate tissue accumulation. Chemotherapeutic agents share one characteristic: they are usually more effective in killing or damaging malignant cells than normal cells. However, the fact that they do harm normal cells indicates their potential for toxicity. Animal tumor investigations and human clinical trials have shown that drug combinations produce higher rates of objective response and longer survival than single agents. Combination drug therapy is, therefore, the basis for most chemotherapy employed at present (DeVita, V. T. et. al., 1995, Cancer 35:98).
Cancer treatment requires inhibitions of a variety of factors including tumor cell proliferation, metastatic dissemination of cancer cells to other parts of the body, invasion, tumor-induced neovascularization, and enhancement of host immunological responses and cytotoxicity. Conventional cancer chemotherapeutic agents have often been selected on the basis of their cytotoxicity to tumor cells. However, some anticancer agents have adverse effects on the patient's immune system. Thus it would be greatly advantageous if a cancer therapy or treatment could be developed that would afford non-cytotoxic protection against factors that might lead to progression of tumors.
Because hormone therapy as well as chemotherapy is effective in controlling advanced breast cancer, it has been used as an adjuvant to mastectomy in primary breast cancer. Patients with ER+ or ER- tumors benefit from adjuvant chemotherapy. However, tamoxifen used alone as an adjuvant to mastectomy for breast cancer shows benefit in extending disease-free and overall survival (Cummings, F. J. et al., 1985, Ann. Intern. Med. 103;324).


The invention relates to wireless communications systems. In particular, the invention relates to initialization of a convolutional decoder that has missed a portion of a continuously encoded symbol stream.
A wireless communication system may comprise multiple remote units and multiple base stations. FIG. 1 exemplifies an embodiment of a terrestrial wireless communication system with three remote units 10A, 10B and 10C and two base stations 12. In FIG. 1, the three remote units are shown as a mobile telephone unit installed in a car 10A, a portable computer remote 10B, and a fixed location unit 10C such as might be found in a wireless local loop or meter reading system. Remote units may be any type of communication unit such as, for example, hand-held personal communication system units, portable data units such as a personal data assistant, or fixed location data units such as meter reading equipment. FIG. 1 shows a forward link 14 from the base station 12 to the remote units 10 and a reverse link 16 from the remote units 10 to the base stations 12.
Communication between remote units and base stations, over the wireless channel, can be accomplished using one of a variety of multiple access techniques which facilitate a large number of users in a limited frequency spectrum. These multiple access techniques include time division multiple access (TDMA), frequency division multiple access (FDMA), and code division multiple access (CDMA). An industry standard for CDMA is set forth in the TIA/EIA Interim Standard entitled xe2x80x9cMobile Stationxe2x80x94Base Station Compatibility Standard for Dual-Mode Wideband Spread Spectrum Cellular Systemxe2x80x9d, TIA/EIA/IS-95, and its progeny (collectively referred to here as IS-95), the contents of which are incorporated by reference herein in their entirety. Additional information concerning a CDMA communication system is disclosed in U.S. Pat. No. 4,901,307, entitled SPREAD SPECTRUM MULTIPLE ACCESS COMMUNICATION SYSTEM USING SATELLITE OR TERRESTRIAL REPEATERS, (the ""307 patent) assigned to the assignee of the present invention and incorporated in its entirety herein by reference.
In the ""307 patent, a multiple access technique is disclosed where a large number of mobile telephone system users, each having a transceiver, communicate through base stations using CDMA spread spectrum communication signals. The CDMA modulation techniques disclosed in the ""307 patent offer many advantages over other modulation techniques used in wireless communication systems such as TDMA and FDMA. For example, CDMA permits the frequency spectrum to be reused multiple times, thereby permitting an increase in system user capacity. Additionally, use of CDMA techniques permits the special problems of the terrestrial channel to be overcome by mitigation of the adverse effects of multipath, e.g. fading, while also exploiting the advantages thereof.
In a wireless communication system, a signal may travel several distinct propagation paths as it propagates between base stations and remote units. The multipath signal generated by the characteristics of the wireless channel presents a challenge to the communication system. One characteristic of a multipath channel is the time spread introduced in a signal that is transmitted through the channel. For example, if an ideal impulse is transmitted over a multipath channel, the received signal appears as a stream of pulses. Another characteristic of the multipath channel is that each path through the channel may cause a different attenuation factor. For example, if an ideal impulse is transmitted over a multipath channel, each pulse of the received stream of pulses generally has a different signal strength than other received pulses. Yet another characteristic of the multipath channel is that each path through the channel may cause a different phase on the signal. For example, if an ideal impulse is transmitted over a multipath channel, each pulse of the received stream of pulses generally has a different phase than other received pulses.
In the wireless channel, the multipath is created by reflection of the signal from obstacles in the environment such as, for example, buildings, trees, cars, and people. Accordingly, the wireless channel is generally a time varying multipath channel due to the relative motion of the structures that create the multipath. For example, if an ideal impulse is transmitted over the time varying multipath channel, the received stream of pulses changes in time delay, attenuation, and phase as a function of the time that the ideal impulse is transmitted.
The multipath characteristics of a channel can affect the signal received by the remote unit and result in, among other things, fading of the signal. Fading is the result of the phasing characteristics of the multipath channel. A fade occurs when multipath vectors add destructively, yielding a received signal that is smaller in amplitude than either individual vector. For example if a sine wave is transmitted through a multipath channel having two paths where the first path has an attenuation factor of X dB, a time delay of xcex4 with a phase shift of "THgr" radians, and the second path has an attenuation factor of X dB, a time delay of xcex4 with a phase shift of "THgr"+xcfx80 radians, no signal is received at the output of the channel because the two signals, being equal amplitude and opposite phase, cancel each other. Thus, fading may have a severe negative effect on the performance of a wireless communication system.
Typically, modern communication systems use coding to improve immunity to interference and wireless channel noise. Additionally, coding may increase system capacity and improve security. Generally, an information signal is first converted into a form suitable for efficient transmission over the wireless channel. Conversion or modulation of the information signal involves varying a parameter of a carrier wave on the basis of the information signal in such a way that the spectrum of the resulting modulated carrier is confined within the channel bandwidth. At a remote unit, the original message signal is replicated from a version of the modulated carrier received following propagation over the wireless channel. Such replication is generally achieved by using an inverse of the modulation process employed by the base station.
The field of data communications is particularly concerned with optimizing data throughput of a transmission system with a limited signal to noise ratio (SNR). The use of error correcting circuitry, such as encoders and decoders, allows system tradeoffs to be made. For example, smaller SNRs or higher data rates may be used with a particular wireless channel which maintains the same bit error rate (BER).
One class of encoders is known as a convolutional encoder. As is well known in the art, a convolutional encoder converts a sequence of input data bits to a codeword based on a convolution of the input sequence with itself or with another signal. Convolutional encoding of data combined with a convolutional decoder is a well known technique for providing error correction coding and decoding of data. One type of convolutional decoder typically used is a Viterbi decoder.
Coding rate, constraint length, and generating polynomials are used to define a convolutional decoder. A coding rate (k/n) corresponds to the number of coding symbols produced (n) for a given number of input bits (k). A constraint length (K) is defined as the length of a shift register used in convolutional encoding of data. Convolutional codes add correlation to an input data sequence by using delay elements (i.e., shift registers) and modulo adders. Taps between the delay elements may terminate at modulo adders forming a desired generating polynomial.
FIG. 2 is a block diagram of a convolutional encoder 20. The encoder 20 shown contains a shift register 22 tapped at various positions 23A through 23N. The shift register taps terminate at one or more of the modulo-2 adders 24 and 25 forming generator functions g0 and g1. Different generating polynomials can be formed by the selection of which taps terminate at the modulo-2 adders.
Bits enter the encoder at its input 26 one at a time. The outputs of the generator functions are the encoded output symbols C0 and C1. Each of the two generator functions g0 and g1 produces an output symbol for each input bit, which corresponds to a code rate of xc2xd. For an encoder with three generator functions, the code rate is ⅓ and a code rate of {fraction (1/n )} requires n generator functions.
A coding rate of xc2xd has become one of the most popular rates, although other code rates may be used. A constraint length of nine (K=9) is typical in convolutional coding schemes. The convolutional encoder can be thought of as a Finite Impulse Response filter with binary coefficients and length Kxe2x88x921. This filter produces a symbol stream with 2Kxe2x88x921 possible states.
A basic principle of the Viterbi algorithm is to take a convolutionally encoded data stream that has been transmitted over a noisy wireless channel and use a finite state machine to efficiently determine the most likely sequence that was transmitted. A K=9 Viterbi decoder can be thought of as a machine that hypothesizes which of each 256 (2(Kxe2x88x921)) possible states the encoder could have been in given the symbols received. The probability that the encoder transitioned from each of those states to the next set of 256 possible encoder states is determined. The probabilities are represented by quantities called metrics, which are proportional to the negative of the logarithm of the probability. The sum of the metrics is therefore equivalent to the reciprocal of the product of the probabilities. Thus, smaller metrics correspond to higher probability events.
There are two types of metrics: state metrics, sometimes called path metrics; and branch metrics. The state metric represents the probability that the received set of symbols leads to the state with which it is associated. The branch metric represents the conditional probability that the transition from one state to another occurred assuming that the starting state was actually the correct state and given the symbol that was actually received.
In a rate {fraction (1/N )} encoder, there are two possible states leading to any other state, each corresponding to the occurrence of a zero or a one in the right-most bit of the convolutional encoder shift register. The decoder decides which is the more likely state by an add-compare-select (ACS) operation. Add refers to adding each state metric at the preceding level to the two branch metrics of the branches for the allowable transitions. Compare refers to comparing the pair of such metric sums for paths entering a state (node) at the given level. Select refers to selecting the lesser of the two and discarding the other. Thus, only the winning branch, i.e., the branch with the highest probability (smallest metric), is preserved at each node, along with the node state metric. If the two quantities being compared are equal, either branch may be selected, for the probability of erroneous selection will be the same in either case.
The Viterbi algorithm is a computationally efficient method of updating the conditional probabilities of the best state and the most probable bit sequence transmitted from the possible 2Kxe2x88x921 states. In order to compute this probability, all 2Kxe2x88x921 states for each bit must be computed. The resulting decision bits from each of these computations is stored as a single bit in a path memory.
A chain-back operation, an inverse of the encoding operation, is performed in which the C decision bits are used to select an output bit, where C is the chainback distance. After many branches the most probable path will be selected with a high degree of certainty. The path memory depth must be sufficiently long to be governed by the signal-to-noise ratio and not the length of the chain-back memory.
Though it is not necessary for analyzing either the code characteristics or the performance of the optimal decoder, it is useful in understanding both to exhibit the code on a trellis diagram. The term xe2x80x9ctrellisxe2x80x9d is a term which describes a tree in which a branch not only bifurcates into two or more branches but also in which two or more branches can merge into one. A trellis diagram is an infinite replication of the state diagram for an encoder. The nodes (states) at one level in the trellis are reached from the node states of the previous level by the transition through one branch, corresponding to one input bit, as determined by the state diagram. Any codeword of a convolutional code corresponds to the symbols along a path (consisting of successive branches) in the trellis diagram.
A simple embodiment of the encoder of FIG. 2 is illustrated in FIG. 3. FIG. 3 illustrates a convolutional encoder with a code rate of xc2xd and a constraint length of 3. As shown in FIG. 3, the convolutional encoder has three taps 31, 32 and 33. The taps terminate at two modulo 2 adders 35 and 36 forming generator functions g0=510 and g1=710. The output of the generator functions become the encoded output symbols C0 and C1, respectively.
FIG. 4 is a trellis diagram showing the possible paths of the convolutional encoder illustrated in FIG. 3. The encoder is assumed to begin in the zero state. Each possible state is represented in the trellis diagram by a node 42. In each state the next input bit into the encoder may be either a zero or a one and a corresponding set of symbols are generated in each generator function. In FIG. 4, input bits 44 at each state are represented on their associated path. The output code symbols C0, indicated as 46, and C1, indicated as 47, generated from the input of each bit are represented in the diagram on the associated path. As illustrated in this simple example, each set of code symbols received at the remote unit is influenced from previously input data bits at the encoder. Thus, in typical operation, a convolutional decoder receives a continuous uninterrupted stream of code symbols with each symbol influenced by the preceding input data.
In a typical CDMA communication system, remote units only sporadically establish bidirectional communication with a base station. For example, a cellular telephone remains idle for significant periods of time when no call is in process. To ensure that any message directed to a remote unit is received, the remote unit must continuously monitor the communication channel even while it is idle. For example, while idle, the remote unit monitors the forward link channel from the base station to detect incoming calls. During such idle periods, the cellular telephone continues to consume power to sustain the elements necessary to monitor for signals from the base stations. Many remote units are portable and are powered by an internal battery. For example, personal communication system (PCS) handsets are almost exclusively battery-powered. The consumption of battery resources by the remote unit in idle mode decreases the battery resources available to the remote unit when a call is placed or received. Therefore, it is desirable to minimize power consumption in a remote unit in the idle state and thereby increase battery life.
One means of reducing remote unit power consumption in a communication system is disclosed in U.S. Pat. No. 5,392,287, entitled APPARATUS AND METHOD FOR REDUCING POWER CONSUMPTION IN A MOBILE COMMUNICATION RECEIVER (the ""287 patent), assigned to the assignee of the present invention and hereby incorporated in its entirety herein by reference. In the ""287 patent, a technique for reducing power consumption in a remote unit operating in an idle mode (i.e. a remote unit which is not engaged in bidirectional communication with a base station) is disclosed. In idle, each remote unit periodically enters an xe2x80x9cactivexe2x80x9d state during which it prepares to and receives messages on a forward link communication channel. In the time period between successive active states, the remote unit enters an xe2x80x9cinactivexe2x80x9d state. During the remote unit""s inactive state, the base station does not send any messages to that remote unit, although it may send messages to other remote units in the system that are in the active state.
As disclosed in the ""287 patent, a base station broadcast messages which are received by all remote units within the base station coverage area on a xe2x80x9cpaging channel.xe2x80x9d All idle remote units within the base station coverage area monitor the paging channel. The paging channel is divided in the time dimension into a continuous stream of xe2x80x9cslots.xe2x80x9d Each remote unit operating in slotted mode monitors only specific slots which have been assigned to it as active (assigned) slots. The paging channel continually transmits convolutional encoded messages in numbered slots, repeating the slot sequence, such as for example, every 640 slots. When a remote unit enters the coverage area of a base station, or if a remote unit is initially powered on, it communicates its presence to a preferred base station. Typically the preferred base station is the base station which has the strongest pilot signal as measured by the remote unit.
The preferred base station, along with a plurality of geographically near neighboring base stations, assign a slot, or a plurality of slots, within their respective paging channels, for the remote unit to monitor. The base station uses the slots in the paging channel to transmit control information to a remote unit, if necessary. The remote unit may also monitor a timing signal from the preferred base station allowing the remote unit to align, in the time dimension, to the base station slot timing. By aligning in the time dimension to the preferred base station slot timing, the remote unit can determine when a paging channel slot sequence begins. Thus, knowing when the paging channel slot sequence begins, which slots are assigned for it to monitor, the total number of slots in the repetitive paging channel sequence of slots, and the period of each slot, the remote unit is able to determine when its assigned slots occur.
Generally, the remote unit is in the inactive state while the base station is transmitting on the paging channel in slots which are not within the remote unit""s assigned set. While in the inactive state, the remote unit does not monitor timing signals transmitted by the base station, maintaining slot timing using an internal clock source. Additionally, while in the inactive state the remote unit may remove power and/or clocks from selected circuitry, such as, for example, circuits which monitor the wireless channel and the decoder. Using its internal timing, the remote unit transits to its active state a short period of time before the next occurrence of an assigned slot.
When transiting to the active state, the remote unit applies power to circuitry that monitors the wireless channel. After the remote unit has reacquired the base station, the remote unit begins receiving a stream of coded symbols and clocks the coded symbols into the decoder. The decoder uses the coded symbols to continue building a trellis stored in the decoder. However, because the stream of coded symbols has been interrupted, the symbol codes being received by the remote unit have no relationship to the symbols that built the trellis stored within the decoder. Therefore the remote unit must receive sufficient code symbols prior to its assigned slot to ensure that proper decoding of the code word is accomplished. For example, the paging channel used in IS-95 is continuously encoded with a constraint length 9 convolutional code. A decoder used to decode the IS-95 paging channel may need to decode 116 data bits to properly initialize its state metrics and insure valid data is output from the decoder.
When the remote unit enters the active state, it may receive messages in its assigned slots in the paging channel and respond to commands from the base station. For example, the remote unit may be commanded to activate a xe2x80x9ctrafficxe2x80x9d channel to establish a bi-directional communication link for conducting subsequent voice communication in response to an incoming call. If there is no message from the base station, or no command requesting the remote unit to remain active, at the end of the assigned slot the remote unit returns to the inactive state. In addition, the remote unit returns to the inactive state immediately if commanded to do so by the base station.
Therefore, there is a need in the art for a method and apparatus to decrease the number of code symbols required to properly decode an interrupted stream of code words.
The invention addresses these and other needs by providing a system and method wherein the convergence of a convolutional decoder is improved. In one aspect of the invention, the remote unit comprising the convolution decoder receives an interrupted stream of code symbols. Prior to decoding the symbols, the state metrics of the trellis residing within the decoder are initialized.
In one aspect of the invention, when the remote unit receives the interrupted stream of code symbols, the pattern of code symbols transmitted just prior to the code symbols received are known. The state metrics of the trellis are then biased towards states which would have been valid only if the previous code symbols would have been received. The remaining invalid states of the trellis are initialized with a high state metric. Therefore, during the decoding process, the decoder is biased toward the valid states.
In another aspect of the invention, the metrics of all states in the trellis residing in the encoder are initialized to 0 or some other constant value. Therefore, there is not a bias towards any particular state within the trellis.


The grid of electric power distribution in the United States includes main, secondary and tertiary high voltage (high tension) cables installed underground in conduits and ducts, normally accessible by manholes. In a large Metropolitan area, such as New York City, servicing millions of customers, there are numerous cables installed in cable bundles in a particular duct or conduit. Under normal operation, the cable jacket and insulation provides adequate thermal and electrical isolation to prevent arcing (electrical discharge) between cables.
High ambient temperatures and low air circulation in manholes, especially during the summer months, combined with the heat generated by the electric cables, can cause electrical breakdown of cable insulation, arcing and cable fires that rapidly transfer fire and heat to adjacent cables, causing further arcing. The result is a chain reaction that causes multiple cable failures interrupting power to all, or nearly all, cables in a manhole, duct or conduit. Damage spread over a wide area results in power outages to many people, requiring days or weeks to repair and restore service.
Additionally, the high levels of smoke and toxic products of combustion will endanger the lives of many people in adjacent areas, such as subway systems, as well as many individuals attempting to enter the affected areas to service or replace damaged cables.
Current cable wraps provide sufficient high voltage protection under normal conditions, but little, if any, fire and heat protection. Many of these wraps were tested and certified more than thirty years ago based on very low standards of fire protection and no requirements for limitations on smoke or toxic products of combustion. The more recently developed wraps continue to use standards for fire protection that do not adequately model the severe conditions in installations today. Further, they do not provide fire, high voltage and environmental resistance necessary for adequate protection in severe conditions.
There is a need for high performance, intumescent and high voltage wrap to isolate each cable under ordinary conditions and more importantly to provide the necessary isolation for cables in the event of overload, dielectric breakdown, and power surge in intense heat that could otherwise lead to a fire. The wrap isolates a cable failure and power can be switched, either by an operator or automatically by the system, to maintain continuity. Customers serviced by the failed cable, or adjacent cables, experience no disruption of service or only momentary disruption due to a switch over.
Intumescent coatings have been known for many years and have been used to provide thermal protection for many substrates including; wood and wood products, metals, fiberglass, and many types of plastics. However, an intumescent coating applied to these aforementioned substances is impractical for the fire and high voltage protection of electrical cables. Currently there is no single fire and high voltage wrap for electrical cables having all the following features:
1) Providing an effective cable wrap that substantially reduces, or eliminates fire spread along cables and heat transmission to adjacent cables.
2) Provide an effective intumescent fire-retardant coating that substantially reduces, or eliminates, smoke and toxic products of combustion from cable insulation when exposed directly or indirectly to a fire.
3) Provide an effective intumescent fire-retardant coating that is noncombustible and will prevent or eliminate ignition of cable insulation.
4) Provide a cable wrap with superior mechanical properties, including flexibility that allows easy handling, and wrapping of electrical cables of varying diameters.
5) Provide a durable and resistant coating to resist abrasion, impact, water, hydrocarbons, chemicals and other environmental factors associated with underground cable installation.
6) Provide a wrap that has high dielectric breakdown strength providing electrical insulation to 50 kVolts.
7) Provide a wrap that has very low ampacity deration, to maintain the high electric current carrying capacity of the cables.
8) Provide a cable wrap that is easily manufactured for mass production.


The present invention relates to a nonvolatile memory device, a method of manufacturing the device, and a method of driving the device, and more particularly, to a nonvolatile memory in which the coupling ratio of the memory cells is increased without increasing cell size, through the structure and operation of a "program assist plate," thereby lowering the operating voltage and increasing the operating speed of the device. The invention may be used in many different types of nonvolatile memory devices, including NAND, NOR, AND, DINOR and other devices.
In a NOR-type electrically erasable programmable read-only memory(EEPROM), two facing memory cells share one bitline contact and one source line, and the memory cells in a row are connected to one bitline. Thus, it is difficult to highly integrate the NOR-type structure, although its high cell current allows it to operate at high speeds.
In a NAND-type structure, two cell strings share one bitline contact and one source line. In one cell string, a plurality of cell transistors are connected in series to the bitline. Accordingly, the NAND-type structure can easily obtain a high level of integration, but it is typically slower than the NOR-type structure due to its low cell current. Because the NAND-type memory cell can be more highly integrated than the NOR-type memory cell, it is generally preferable to employ the NAND-type memory structure for increasing the capacity of a memory device. However, this invention is not limited to application in only NAND type devices
The EEPROM NAND string structure and the basic operation of the NAND-type EEPROM are described below, referring to the accompanying drawings.
FIG. 1 is a plan view showing the layout with respect to one string in a typical NAND-type nonvolatile memory device, and FIG. 2 is an equivalent circuit diagram of the structure shown in FIG. 1.
Referring to FIGS. 1 and 2, each string of a NAND-type nonvolatile memory device is formed by sequentially connecting a string selection transistor S1, a plurality of cell transistors C1, . . . , Cn and a source selection transistor S2 in series between the bitline B/L and a source line S/L in an area represented by a width x and a length y.
FIG. 3A is a plan view of a transistor cell used in forming each string of the nonvolatile memory device, and FIG. 3B is a sectional view taken along line I-I' of FIG. 3A.
In FIG. 3A, reference numeral 26 indicates a mask pattern for forming an active region, reference numeral 24 indicates a mask pattern for forming a control gate, and reference numeral 22 indicates a mask pattern for forming a floating gate.
Referring to FIG. 3B, each transistor cell C1, . . . , Cn of FIG. 1 in the string consists of a floating gate 32, a control gate 34 and a N-type source/drain 36, which are sequentially deposited on a P-type semiconductor substrate 30, with an interdielectric layer inserted therebetween. The programming, erasing and reading of a NAND-type nonvolatile memory device having this structure is described below.
The NAND-type nonvolatile memory is programmed by tunneling an electric charge from a channel region of the cell transistor to the floating gate thereof, to thereby store information. For example, if information is to be programmed or stored in the first transistor cell C1, power supply voltage Vcc is applied to the gate of string select transistor S1, thereby turning on string select transistor S1, and 0V is applied to the gate of source select transistor S2, thereby turning off source select transistor. With reference to FIG. 3B, a programming voltage Vpgm is applied to the control gate 34 of the first transistor cell C1, to thereby generate tunneling. Accordingly, an electric charge in the channel region of the substrate 30 moves to the floating gate 32, to thereby change the threshold voltage Vth of the first transistor cell C1.
After programming, transistor cell C1 will have (approximately) one of two different threshold voltages depending on the charge transferred to the floating gate 32. The first and second threshold voltages may correspond to either a "1" or "0" in a two-state memory device. In a multi-state memory device more than two threshold voltages may be used, thereby storing more than one bit per cell.
A read operation is used to determine the programmed state of the NAND memory cell. For example, referring to FIG. 2, when reading information stored in the first cell transistor C1, the bitline B/L is precharged with a predetermined voltage between approximately 1 V.about.Vcc. Then, Vcc is applied to each control gate of the string select transistor S1, the source select transistor S2 and unselected cell transistors C2, . . . , Cn, (i.e., each cell transistor except for C1) to thereby turn-on the transistors. Approximately 0 V is applied to the control gate of the selected first cell transistor C1, which is between a first threshold voltage of approximately -3 V when a "1" is stored in the cell and a second threshold voltage of approximately 1V when a "0" is stored in cell C1. Thus, if the first cell transistor C1 is turned on, and a current is sensed between the bitline B/L and source line S/L, the state of the first cell transistor C1 is determined as "1". However, if the first cell transistor C1 is turned off, and no (or very little) current is sensed between the bitline B/L and the source line S/L, the state of the first cell transistor C1 is determined as "0". Alternatively, no current could correspond to a "1" and a sensed current could correspond to a "0".
The erasing operation is performed by tunneling an electric charge from the floating gate 32 to the channel region of the substrate 30 (FIG. 3B), thereby erasing information stored in the cell. For example, referring to FIG. 2, when information is to be erased from the first cell transistor C1, the cell string is placed in a floating state by disconnecting it from the bitline B/L and the source line S/L by turning off the string select transistor S1 and source select transistor S2. A voltage of 0V is applied to all 25 wordlines of a selected block of memory cells C1, C2, . . . Cn. Further, referring to FIG. 3B, an erase voltage Verase is applied to the substrate 30, thereby generating tunneling from the floating gate 32 to the substrate 30. Thus, the electric charge on the floating gate 32 is moved to the substrate 30, thereby changing the threshold voltage of the selected memory cells.
In the operation of the nonvolatile memory device described above, a high-voltage of approximately 20V is required to program or erase the memory cells by Fowler-Nordheim (referred to as "F-N") tunneling. A charge pumping circuit is required to supply a high voltage for programming and erasing, which results in increased chip size and power consumption. Accordingly, in order to increase the density of a nonvolatile memory device, it is important to increase the efficiency of both erasing and programming, and thereby lower the power requirements for Vpgm and Verase.
In order to enhance the operating characteristics without lowering the reliability of the nonvolatile memory device, the capacitance of the structure corresponding to the interdielectric layer deposited between the control gate 34 and the floating gate 32 must be increased, and the program/erase voltage must be lowered. The capacitance may be increased by reducing the thickness of the interdielectric layer or increasing the contact area of the control gate 34 and the floating gate 32. If the capacitance is increased by reducing the thickness of the interdielectric layer, the data retention capability of the nonvolatile memory device is reduced, and the insulation of the interdielectric layer may be broken during programming and erasing. In addition, the process for producing an interdielectric layer of reduced thickness is difficult. However, a method has recently been developed for increasing the contact area between the control gate 34 and the floating gate 32.
FIG. 4 is a plan view showing a layout of a conventional NAND-type nonvolatile memory device, disclosed in IEDM Tech. Dig. 1994, pp. 61-64, which is incorporated by reference herein. This article discloses a structure and method for obtaining high-integration and increased capacitance with respect to the interdielectric layer by increasing the effective surface area.
In FIG. 4, reference numeral 40 denotes a mask pattern for defining an active region, reference numeral 42 denotes a mask pattern for forming a floating gate, reference numeral 44 denotes a mask pattern for forming a control gate, and reference numeral 46 denotes a mask pattern for forming a bitline contact. The mask pattern 42 for forming the floating gate completely overlaps with the mask pattern 40 for defining an active region. That is, the floating gate is self-aligned on the active region, which leads to high-integration.
FIG. 5 is a sectional view taken along line II-II' of FIG. 4, where reference numeral 50 denotes a semiconductor substrate, reference numeral 52 denotes a floating gate, reference numeral 54 denotes an interdielectric layer, reference numeral 56 denotes a control gate, and reference numeral 58 denotes an isolation film. In this nonvolatile memory device, the area of the interdielectric layer 54 between the floating gate 52 and the control gate 56 is determined only by the width of the active region of the cell transistor. The active regions are the portions of the substrate 50 located between adjacent isolation films 58. Note that the floating gate 52 is not formed on the isolation film 58. Accordingly, the capacitance related to the interdielectric layer 54 is lowered, and the resulting device requires a high voltage for programming and erasing.
To solve the above problems, the thickness of the floating gate 52 of FIG. 5 is increased. However, the thick floating gate 52 structure causes two problems. First, when the control gate 56, the interdielectric layer 54, and the floating gate 52 are patterned in accordance with this process, a vertically high interdielectric layer 54 must be formed on the sidewalls of the thick floating gate 52. In addition, it is difficult to etch the thick floating gate 52 in the source/drain region (not shown) of a cell transistor.


The plasma display technology is relatively new and has generated a whole new set of requirements for regulated power supplies. A plasma display will typically require a power supply which generates highly regulated 200 volt D.C. and 5 volt D.C. outputs. The supply voltages are generally isolated from other voltages within the computer and the power supply must generally be current limited and short circuit protected. An additional requirement of the power supply used with a plasma display is that the 200 volt D.C. output must be enabled only after other computer operational voltages have stabilized. If other operational voltages fail for any reason, the 200 volt D.C. voltage for the plasma display must be disabled. This sequencing of the 200 volt D.C. output is required to prevent damage to the plasma display.
An adjustable three terminal regulator, due to its not having a ground end, can regulate high voltages as long as its differential voltage rating is not exceeded. In Linear Technology Data Book, 1986, at pages 71-72, a circuit is described which will protect the regulator from a short circuit condition but, as stated in the reference, such a scheme for high voltage regulation is limited by the power dissipation capabilities of the device in series with the regulator.
With an input voltage of 250 volts and a typical current of 200 milliamps, the power dissipated is approximately 50 watts. Should a short circuit condition exist for an extended period, which may occasionally occur, an excessive amount of heat is generated requiring an unacceptably large power transistor and heat sink.
An attempt to reduce the power dissipation by causing the current limit circuitry to oscillate causes the load change to be reflected to other outputs, causing high ripple content at a frequency approximating the current limit oscillations.
Further, providing an input voltage to the regulator that will not exceed the safe operating levels of the regulator becomes a concern due to the voltage overshoot caused by transformer leakage inductance. Left unimpeded, this inductance can allow the input voltage to vary by as much as 30%.
The plasma display technology further requires a sequencing or timing control of the 200 volt D.C. output to the display to prevent possible damage o the display. The display voltages must be enabled only after all other computer operational voltages have been established and have stabilized at their normal level. In the event one of those operational voltages fail, the display voltages must be disabled. This sequencing or timing requirement requires coordination of the display voltages with other operational voltages within the computer.


From DE 10 2007 056 638 A1 a generic device is known for assembling a composite arrangement, consisting of a plurality of functional elements having an aperture for a shaft, in particular cams, balancing masses, toothed wheels and/or bearings, in a predetermined angular position on the shaft, wherein the device has a plurality of retaining devices intended respectively for a functional element.
From DE 10 2008 064 194 A1 an apparatus is known for the positioning of a plurality of functional elements having an aperture for a shaft, in particular cams, in a predetermined angular position on the shaft, wherein the apparatus has a plurality of mounts intended respectively for a functional element, which mounts are equipped respectively with a moulding fixing the angular position of the respective functional element corresponding to the angular position on the shaft. The mounts are able to be positioned here such that the apertures of the functional elements lie on a shared horizontal line.
From DE 10 2009 060 350 A1 an apparatus is known for assembling a shaft carrying functional elements, wherein the apparatus comprises a machine platform, on which a plurality of positioning discs for the aligned, correct positioning of the functional elements is arranged in such a manner that a shaft can be pushed in. The positioning discs are reversibly fixed on a frame, which in turn is reversibly fixed on the machine platform. This is intended to make possible rapid changing of a production process by having several frames available.


The present invention relates to a pendulum mechanism, adapted to be attached to the casing of a clockwork for use together with the clockwork, which operates independently of the clockwork, and which comprises a pivot bearing, a pendulum arm mounted on the bearing for swinging motion, and an electrical drive mechanism for the pendulum comprising a permanent magnet carried by the pendulum adjacent its lower end and cooperating with a coil mounted on a circuit board located below the clockwork casing and carrying the electronics for driving the pendulum.
An autonomously operating pendulum arrangement, which can be attached to a crystal-controlled clockwork, is discussed in Haag et al U.S. Pat. No. 4,043,118, assigned to the assignee of the instant application. This known arrangement consists of a pendulum housing adapted to be fastened to the lower side of the clockwork casing and carrying a circuit board on which are mounted the electrical components required to drive the pendulum. The arrangement includes a bearing device for the pendulum arm which bearing device is attached as a separate element to the upper side of the clockwork casing, and a device for suspending the pendulum is provided at the lower side of the pendulum arm. The housing is also provided with a lateral arm on which the bell of a clock-striking system can be mounted.
This known arrangement has the disadvantage that the proper functioning of its various components, i.e., the overall pendulum arrangement as well as the associated chime system, can be checked out for proper operation only when all of the separate parts of the pendulum have been assembled on the clockwork casing. There is, in addition, the disadvantage that the pendulum arm is substantially U-shaped and envelopes the clockwork laterally, and this requires a comparatively large amount of space for the swinging motion of the pendulum. The known arrangement, moreover, cannot be used in conjunction with comparatively small pendulum clocks, for example table clocks; and the lateral placement of the bell of the clock-striking system precludes use of the arrangement in small-sized classic clocks.
The present invention is intended to obviate these disadvantages of the prior art, and is concerned with a novel autonomous pendulum mechanism which can be connected to the casing of the clockwork in the form of a completely separate assembled structural unit whose functioning can be checked out independently of the clockwork. Moreover, the unitary pendulum mechanism of the present invention is so arranged that a chiming system can be incorporated therein as part of the pendulum unit, so that its proper operation can also be checked out independently of the clockwork.


In a communication system, a light modulator may amplitude modulate a light source based on a data signal, and the resulting modulated light, which conveys the data signal, is transmitted to a distant light receiver. In some situations it is advantageous to be able to transmit multiple data signals with different data rates, simultaneously. A conventional light modulator is limited in that it is not capable of simultaneously amplitude modulating a light source with such data signals having the different data rates. Instead, multiple conventional light modulators must be operated in parallel each to transmit a different one of the data signals. This wastes power and increases costs.
In the drawings, the leftmost digit(s) of a reference number identifies the drawing in which the reference number first appears.


The allenyl .beta.-lactam compound represented by the general formula (4) given below is conventionally prepared, for example, by reacting a tertiary organic base with the starting material in an organic solvent according to the process disclosed in JP-A-282359/1992. However, this compound is unstable when present in the resulting reaction mixture, which therefore usually requires cumbersome repeated procedures for extraction and concentration after the completion of the reaction. These procedures require time in the case of a large scale production, consequently entailing problems such as a marked reduction in the yield of the isolated product. Thus, processes still remain to be developed which are satisfactorily feasible for the preparation of allenyl .beta.-lactam compounds.
Reports have been made on widely acceptable processes for preparing 3-halogenated cephem derivatives represented by the general formula (6) given below. These processes include a process which uses a compound of the general formula (7), i.e., 3-hydroxycephem compound, as the starting material and involves the conversion of hydroxyl group to trifluoromesyloxy group first and the subsequent reaction with a lithium halide as described in J. Org. Chem., 54, 4962(1989), a process wherein a reactive chlorine or bromine compound (such as phosphorus trichloride, phosphorus oxychloride or thionyl bromide) is reacted with a 3-- hydroxycephem compound in dimethylformamide as disclosed in JP-A-116095/1974), and further a process wherein an alkali metal salt or alkaline-earth metal salt of a halogen is reacted with an allenyl .beta.-lactam compound as disclosed in JP-A-282387/1992 ##STR1## wherein R.sup.1, R.sup.2 and R.sup.3 are as defined below.
The first of the processes requires the use of the 3-hydroxycephem compound as the starting material which compound itself is difficult to prepare, so that the process is in no way practically feasible. The second process inevitably forms 3-sulfonylcephem or 3-thiocephem as a by-product due to the recombination of sulfinate ion or thiolate ion which is released on ring closure, consequently giving the desired 3-halogenated cephem derivative in a yield of as low as up to 70%.
An object of the present invention is to provide a process wherein a .beta.-lactam compound represented by the general formula (1) is used as the starting material for preparing an allenyl .beta.-lactam compound of the general formula (4) as isolated with stability and high purity in a high yield by a simplified procedure.
Another object of the invention is to overcome the drawbacks of the conventional production processes described and to provide a widely-useful process for preparing the desired 3-halogenated cephem derivative in a high yield with a high purity.


The use of computing devices, such as cellular phones and personal digital assistants (PDAs) has grown rapidly. Such devices provide many different functions to users through different types of interfaces, such as keypads and displays. Some computing devices utilize motion as an interface by detecting tilt of the device by a user. Some implementations of a motion interface involve tethering a computing device with fishing lines or carrying large magnetic tracking units that require large amounts of power.


Certain embodiments of the present invention are directed to integrated circuits. More particularly, some embodiments of the invention provide a system and method for stage-based control related to TRIAC dimmer. Merely by way of example, some embodiments of the invention have been applied to driving one or more light emitting diodes (LEDs). But it would be recognized that the invention has a much broader range of applicability.
A conventional lighting system may include or may not include a TRIAC dimmer that is a dimmer including a Triode for Alternating Current (TRIAC). For example, the TRIAC dimmer is either a leading-edge TRIAC dimmer or a trailing-edge TRIAC dimmer. Often, the leading-edge TRIAC dimmer and the trailing-edge TRIAC dimmer are configured to receive an alternating-current (AC) input voltage, process the AC input voltage by clipping part of the waveform of the AC input voltage, and generate a voltage that is then received by a rectifier (e.g., a full wave rectifying bridge) in order to generate a rectified output voltage.
FIG. 1 shows certain conventional timing diagrams for a leading-edge TRIAC dimmer and a trailing-edge TRIAC dimmer. The waveforms 110, 120, and 130 are merely examples. Each of the waveforms 110, 120, and 130 represents a rectified output voltage as a function of time that is generated by a rectifier. For the waveform 110, the rectifier receives an AC input voltage without any processing by a TRIAC dimmer. For the waveform 120, an AC input voltage is received by a leading-edge TRIAC dimmer, and the voltage generated by the leading-edge TRIAC dimmer is received by the rectifier, which then generates the rectified output voltage. For the waveform 130, an AC input voltage is received by a trailing-edge TRIAC dimmer, and the voltage generated by the trailing-edge TRIAC dimmer is received by the rectifier, which then generates the rectified output voltage.
As shown by the waveform 110, each cycle of the rectified output voltage has, for example, a phase angel (e.g., ϕ) that changes from 0° to 180° and then from 180° to 360°. As shown by the waveform 120, the leading-edge TRIAC dimmer usually processes the AC input voltage by clipping part of the waveform that corresponds to the phase angel starting at 0° or starting at 180°. As shown by the waveform 130, the trailing-edge TRIAC dimmer often processes the AC input voltage by clipping part of the waveform that corresponds to the phase angel ending at 180° or ending at 360°.
Various conventional technologies have been used to detect whether or not a TRIAC dimmer has been included in a lighting system, and if a TRIAC dimmer is detected to be included in the lighting system, whether the TRIAC dimmer is a leading-edge TRIAC dimmer or a trailing-edge TRIAC dimmer. In one conventional technology, a rectified output voltage generated by a rectifier is compared with a threshold voltage Vth_on in order to determine a turn-on time period Ton. If the turn-on time period Ton is approximately equal to the duration of a half cycle of the AC input voltage, no TRIAC dimmer is determined to be included in the lighting system; if the turn-on time period Ton is not approximately equal to but is smaller than the duration of a half cycle of the AC input voltage, a TRIAC dimmer is determined to be included in the lighting system. If a TRIAC dimmer is determined to be included in the lighting system, a turn-on voltage slope Von_slope is compared with the threshold voltage slope Vth_slope. If the turn-on voltage slope Von_slope is larger than the threshold voltage slope Vth_slope, the TRIAC dimmer is determined to be a leading-edge TRIAC dimmer; if the turn-on voltage slope Von_slope is smaller than the threshold voltage slope Vth_slope, the TRIAC dimmer is determined to be a trailing-edge TRIAC dimmer.
If a conventional lighting system includes a TRIAC dimmer and light emitting diodes (LEDs), the light emitting diodes may flicker if the current that flows through the TRIAC dimmer falls below a holding current that is, for example, required by the TRIAC dimmer. As an example, if the current that flows through the TRIAC dimmer falls below the holding current, the TRIAC dimmer may turn on and off repeatedly, thus causing the LEDs to flicker. As another example, the various TRIAC dimmers made by different manufacturers have different holding currents ranging from 5 mA to 50 mA.
The light emitting diodes (LEDs) are gradually replacing incandescent lamps and becoming major lighting sources. The LEDs can provide high energy efficiency and long lifetime. The dimming control of LEDs, however, faces significant challenges because of insufficient dimmer compatibility. For certain historical reasons, the TRIAC dimmers often are designed primarily suitable for incandescent lamps, which usually include resistive loads with low lighting efficiency. Such low lighting efficiency of the resistive loads often helps to satisfy the holding-current requirements of TRIAC dimmers. Hence the TRIAC dimmers may work well with the incandescent lamps. In contrast, for highly efficient LEDs, the holding-current requirements of TRIAC dimmers usually are difficult to meet. The LEDs often need less amount of input power than the incandescent lamps for the same level of illumination.
In order to meet the holding-current requirements of the TRIAC dimmers, some conventional techniques use a bleeder for a lighting system. FIG. 2 is a simplified diagram of a conventional lighting system that includes a bleeder. As shown, the conventional lighting system 200 includes a TRIAC dimmer 210, a rectifier 220, a bleeder 224, a diode 226, capacitors 230, 232, 234, 236 and 238, a pulse-width-modulation (PWM) controller 240, a winding 260, a transistor 262, resistors 270, 272, 274, 276, 278 and 279, and one or more LEDs 250. The PWM controller 240 includes controller terminals 242, 244, 246, 248, 252, 254, 256 and 258. For example, the PWM controller 240 is a chip, and each of the controller terminals 242, 244, 246, 248, 252, 254, 256 and 258 is a pin. In yet another example, the winding 260 includes winding terminals 263 and 265.
The TRIAC dimmer 210 receives an AC input voltage 214 (e.g., VAC) and generates a voltage 212. The voltage 212 is received by the rectifier 220 (e.g., a full wave rectifying bridge), which then generates a rectified output voltage 222. The rectified output voltage 222 is larger than or equal to zero. The resistor 279 includes resistor terminals 235 and 239, and the capacitor 236 includes capacitor terminals 281 and 283. The resistor terminal 235 receives the rectified output voltage 222. The resistor terminal 239 is connected to the capacitor terminal 281, the controller terminal 252, and a gate terminal of the transistor 262. The gate terminal of the transistor 262 receives a gate voltage 237 from the resistor terminal 239, the capacitor terminal 281, and the controller terminal 252. The capacitor terminal 283 receives a ground voltage.
As shown in FIG. 2, the rectified output voltage 222 is used to charge the capacitor 236 through the resistor 279 to raise the gate voltage 237. In response, if the result of the gate voltage 237 minus a source voltage at a source terminal of the transistor 262 reaches or exceeds a transistor threshold voltage, the transistor 262 is turned on. When the transistor 262 is turned on, through the transistor 262 and the controller terminal 254, a current flows into the PWM controller 240 and uses an internal path to charge the capacitor 232. In response, the capacitor 232 generates a capacitor voltage 233, which is received by the controller terminal 244. If the capacitor voltage 233 reaches or exceeds an undervoltage-lockout threshold of the PWM controller 240, the PWM controller 240 starts up.
After the PWM controller 240 has started up, a pulse-width-modulation (PWM) signal 255 is generated. The PWM signal 255 has a signal frequency and a duty cycle. The PWM signal 255 is received by the source terminal of the transistor 262 through the terminal 254. The transistor 262 is turned on and off, in order to make an output current 266 constant and provide the output current 266 to the one or more LEDs 250, by working with at least the capacitor 238.
As shown in FIG. 2, a drain voltage at a drain terminal of the transistor 262 is received by a voltage divider that includes the resistors 276 and 278. The drain terminal of the transistor 262 is connected to the winding terminal 265 of the winding 260, and the winding terminal 263 of the winding 260 is connected to the capacitor 230 and the resistor 279. In response, the voltage divider generates a voltage 277, which is received by the controller terminal 256. The PWM controller 240 uses the voltage 277 to detect the end of a demagnetization process of the winding 260. The detection of the end of the demagnetization process is used to control an internal error amplifier of the PWM controller 240, and through the controller terminal 246, to control charging and discharging of the capacitor 234.
Also, after the PWM controller 240 has started up, the resistor 274 is used to detect a current 261, which flows through the winding 260. The current 261 flows from the winding 260 through the resistor 274, which in response generates a sensing voltage 275. The sensing voltage 275 is received by the PWM controller 240 at the controller terminal 258, and is processed by the PWM controller 240 on a cycle-by-cycle basis. The peak magnitude of the sensing voltage 275 is sampled, and the sampled signal is sent to an input terminal of the internal error amplifier of the PWM controller 240. The other input terminal of the internal error amplifier receives a reference voltage Vref.
As shown in FIG. 2, the rectified output voltage 222 is received by a voltage divider that includes the resistors 270 and 272. In response, the voltage divider generates a voltage 271, which is received by the controller terminal 242. The PWM controller 240 processes the voltage 271 and determines phase angle of the voltage 271. Based on the detected range of phase angle of the voltage 271, the PWM controller 240 adjusts the reference voltage Vref, which is received by the internal error amplifier.
The bleeder 224 is used to ensure that, when the TRIAC dimmer 210 is fired on, an input current 264 that flows through the TRIAC dimmer 210 is larger than a holding current required by the TRIAC dimmer 210, in order to avoid misfire of the TRIAC dimmer 210 and also avoid flickering of the one or more LEDs 250. For example, the bleeder 224 includes a resistor, which receives the rectified output voltage 222 at one resistor terminal of the resistor and receives the ground voltage at the other resistor terminal of the resistor. The resistor of the bleeder 224 allows a bleeder current 268 to flow through as at least part of the input current 264. In another example, if the holding current required by the TRIAC dimmer 210 is small and if the average current that flows through the transistor 262 can satisfy the holding current requirement of the TRIAC dimmer 210, the bleeder 224 is not activated or is simply removed.
As shown in FIG. 2, the lighting system 200 includes, for example, a quasi-resonant system with a buck-boost topology. The output current 266 of the quasi-resonant system is received by the one or more LEDs 250 and is determined as follows:
                              I          o                =                              1            2                    ×                                    V              ref                                      R              cs                                                          (                  Equation                                                              1                )            where I0 represents the output current 266 of the quasi-resonant system of the lighting system 200. Additionally, Vref represents the reference voltage received by the internal error amplifier of the PWM controller 240. Moreover, Rcs represents the resistance of the resistor 274.
FIG. 3 is a simplified diagram showing certain conventional components of the lighting system 200 as shown in FIG. 2. The pulse-width-modulation (PWM) controller 240 includes a dimming control component 300 and a transistor 350. The dimming control component 300 includes a phase detector 310, a reference voltage generator 320, a pulse-width-modulation (PWM) signal generator 330, and a driver 340.
FIG. 4 shows certain conventional timing diagrams for the lighting system 200 as shown in FIGS. 2 and 3. The waveform 471 represents the voltage 271 as a function of time, the waveform 412 represents the phase signal 312 as a function of time, the waveform 475 represents the sensing voltage 275 as a function of time, and the waveform 464 represents cycle-by-cycle average of the input current 264 as a function of time.
As shown by FIGS. 3 and 4, the lighting system 200 uses a closed loop to perform dimming control. The phase detector 310 receives the voltage 271 through the terminal 242, detects phase angle of the voltage 271, and generates a phase signal 312 that indicates the detected range of phase angle of the voltage 271. As shown by the waveform 471, the voltage 271 becomes larger than a dim-on threshold voltage (e.g., Vth_dimon) at time ta and becomes smaller than a dim-off threshold voltage (e.g., Vth_dimon) at time tb. The dim-on threshold voltage (e.g., Vth_dimon) is equal to or different from the dim-off threshold voltage (e.g., Vth_dimoff). The time duration from time ta to time tb is represented by TR, during which the phase signal 312 is at the logic high level, as shown by the waveform 412. The time duration TR represents the detected range of phase angle of the voltage 271.
During the time duration TR, the sensing voltage 275 ramps up and down. For example, during the time duration TR, within a switching period (e.g., TSW), the sensing voltage 275 ramps up, ramps down, and then remains constant (e.g., remains equal to zero) until the end of the switching period (e.g., until the end of TSW).
The phase signal 312 is received by the reference voltage generator 320, which uses the detected range of phase angle of the voltage 271 to generate the reference voltage 322 (e.g., Vref). As shown in FIG. 3, the reference voltage 322 (e.g., Vref) is received by the PWM signal generator 330. For example, the PWM signal generator 330 includes the internal error amplifier of the PWM controller 240. In another example, the PWM signal generator 330 also receives the sensing voltage 275 and generates a pulse-width-modulation (PWM) signal 332. The PWM signal 332 is received by the driver 340, which in response generates a drive signal 342 and outputs the drive signal 342 to the transistor 350. The transistor 350 includes a gate terminal, a drain terminal, and a source terminal. The gate terminal of the transistor 350 receives the drive signal 342. The drain terminal of the transistor 350 is coupled to the controller terminal 254, and the source terminal of the transistor 350 is coupled to the controller terminal 258.
As shown by the waveform 475, the reference voltage 322 (e.g., Vref) is used by the PWM signal generator 330 to generate the PWM signal 332, which is then used to control the peak magnitude (e.g., CS_peak) of the sensing voltage 275 for each PWM cycle during the time duration TR. For example, each PWM cycle corresponds to a time duration that is equal to the switching period (e.g., TSW) in magnitude. In another example, if the detected range of phase angle of the voltage 271 (e.g., corresponding to TR) becomes larger, the reference voltage 322 (e.g., Vref) also becomes larger. In yet another example, if the detected range of phase angle of the voltage 271 (e.g., corresponding to TR) becomes smaller, the reference voltage 322 (e.g., Vref) also becomes smaller.
According to Equation 1, if the reference voltage 322 (e.g., Vref) becomes larger, the output current 266 (e.g., Io) of the quasi-resonant system of the lighting system 200 also becomes larger; if the reference voltage 322 (e.g., Vref) becomes smaller, the output current 266 (e.g., Io) of the quasi-resonant system of the lighting system 200 also becomes smaller.
As shown by FIG. 2, the cycle-by-cycle average of the input current 264 is approximately equal to the sum of cycle-by-cycle average of the output current 266 (e.g., Io) and the bleeder current 268. During the time duration TR, within each switching cycle of the PWM signal 332, the output current 266 changes with time, so the average of the output current 266 within each switching cycle is used to determine the cycle-by-cycle average (e.g., I_PWM_av) of the output current 266 as a function of time. When the time duration TR becomes smaller, the reference voltage 322 (e.g., Vref) also becomes smaller and the one or more LEDs 250 are expected to become dimmer. When the time duration TR becomes too small, the reference voltage 322 (e.g., Vref) also becomes too small and the cycle-by-cycle average (e.g., I_PWM_av) of the output current 266 during the time duration TR becomes smaller than the holding current (e.g., I_holding) required by the TRIAC dimmer 210. In order to avoid misfire of the TRIAC dimmer 210 and also avoid flickering of the one or more LEDs 250, the bleeder current 268 (e.g., I_bleed) is provided in order to increase the cycle-by-cycle average of the input current 264 during the time duration TR. As shown by the waveform 464, the cycle-by-cycle average of the input current 264 during the time duration TR becomes larger than the holding current required by the TRIAC dimmer 210.
As shown in FIG. 3, the driver 340 outputs the drive signal 342 to the transistor 350. The transistor 350 is turned on if the drive signal 342 is at a logic high level, and the transistor 350 is turned off if the drive signal 342 is at a logic low level. When the transistor 262 and the transistor 350 are turned on, the current 261 flows through the winding 260, the transistor 262, the controller terminal 254, the transistor 350, the controller terminal 258, and the resistor 274. If the transistor 350 becomes turned off when the transistor 262 is still turned on, the transistor 262 then also becomes turned off and the winding 260 starts to discharge. If the transistor 350 becomes turned on when the transistor 262 is still turned off, the transistor 262 then also becomes turned on and the winding 260 starts to charge.
As shown in FIGS. 2-4, the lighting system 200 uses a closed loop to perform dimming control. For example, the lighting system 200 detects the range of phase angle of the voltage 271, and based on the detected range of phase angle, adjusts the reference voltage Vref that is received by the internal error amplifier of the PWM controller 240. In another example, the lighting system 200 provides energy to the one or more LEDs 250 throughout the entire time period of each switching cycle during the time duration TR, which corresponds to the unclipped part of the waveform of the AC input voltage 214 (e.g., VAC).
As discussed above, a bleeder (e.g., the bleeder 224) can help a lighting system (e.g., the lighting system 200) to meet the holding-current requirement of a TRIAC dimmer (e.g., the TRIAC dimmer 210) in order to avoid misfire of the TRIAC dimmer (e.g., the TRIAC dimmer 210) and avoid flickering of one or more LEDs (e.g., the one or more LEDs 250). But the bleeder (e.g., the bleeder 224) usually increases heat generation and reduces energy efficiency of the lighting system (e.g., the lighting system 200). Such reduction in energy efficiency usually becomes more severe if a bleeder current (e.g., the bleeder current 268) becomes larger. This reduced energy efficiency often prevents the lighting system (e.g., the lighting system 200) from taking full advantage of high energy efficiency and long lifetime of the one or more LEDs (e.g., the one or more LEDs 250).
Hence it is highly desirable to improve the techniques of dimming control.


1. Field of the Invention
The present invention relates to a method and an apparatus in which a bias of a focus servo circuit is adjusted so that a signal surface of an optical disc may be placed within a focus depth of a laser beam.
2. Description of the Prior Art
Conventionally, after irradiating a laser beam onto a signal surface of an optical disc, an optical disc player converts light reflected therefrom into an electric signal, and thereby reads out data which is recorded on the optical disc.
In the optical disc player known heretofore, it is customary to execute the following servo control.
For example, a light beam emitted from a light source is irradiated to the surface of an optical disc, and a return light beam from the optical disc is received by a photo detector.
A focus error signal and a tracking error signal are detected on the basis of the outputs of focus sensors which are divisions of the photo detector.
The focus error signal and the tracking error signal thus detected are supplied to a focus servo circuit and a tracking servo circuit, respectively and thereby execute focus servo control and tracking servo control.
FIG. 1 is a block diagram for showing a circuit configuration of a conventional optical disc player. As shown in FIG. 1, an optical disc player 1 comprises an optical pickup 2, a tracking servo circuit 5 and a focus servo circuit 6 which receive, via amplifier 3 and 4 respectively, a tracking error signal and a focus error signal obtained on the basis of the difference between the light quantities of individual light receiving elements of a photo detector in optical pickup 2.
Optical disc player 1 further comprises a tracking driver 7 for driving an actuator of optical pickup 2 under control on the basis of a servo signal from tracking servo circuit 5 to thereby move an objective lens in the tracking direction, a focus driver 8 for driving the actuator of optical pickup 2 under control on the basis of a servo signal from focus servo circuit 6 to thereby move the objective lens in the focusing direction, and an adding circuit 9A for applying a focus bias, which is obtained from the focus bias generator circuit 9, to the focus error signal outputted from amplifier 4.
Optical pickup 2 has a known structure wherein an objective lens (not shown) is held to be movable biaxially, so that when a tracking coil and a focus coil provided in the actuator are fed with current, the objective lens can be driven biaxially under control in both of the tracking and focusing directions.
Tracking servo circuit 6 is applied with the tracking error signal obtained from optical pickup 2 and amplified by amplifier 3, and then outputs a tracking control signal to driver 7 in accordance with the tracking error signal so as to minimize the tracking error.
Focus servo circuit 6 is supplied with the focus error signal obtained from optical pickup 2 and amplified by amplifier 4, and then outputs a focus control signal to driver 8 in accordance with the focus error signal so as to minimize the focus error.
Driver 7 serves to drive the actuator of optical pickup 2 in response to the tracking control signal received from tracking servo circuit 5 and thereby moves the objective lens in the tracking direction to minimize the tracking error.
Meanwhile, driver 8 serves to drive the actuator of optical pickup 2 in response to the focus control signal received from focus servo circuit 6 and thereby moves the objective lens in the focusing direction to minimize the focus error.
Since the minimum point of the focus error signal may sometimes fail to coincide with the least jitter point of a reproduced signal, a focus bias obtained from the focus bias generator circuit 9 is applied to the focus error signal for causing the minimum point of the focus error signal to coincide with the least jitter point of a reproduced signal.
In optical disc player 1 of the structure mentioned above, the focus error bias is adjusted in the following manner.
At the time of assembling optical disc player 1, focusing is performed in an on-state of the focus servo, and an adjusting rheostat 9B incorporated in the focus bias generator circuit 9 is manually operated while observing the RF signal from optical pickup 2 and monitoring the value of the jitter, whereby an optimal focus bias is determined with respect to the individual optical disc player 1.
However, in the optical disc player of the above structure, a time required for determining the focus bias is long with another disadvantage of necessitating the adjusting rheostat 9B, hence raising a problem of higher cost with regard to the component parts.
Further, in an operation of recording data on and/or reproducing the same from the optical disc, the refractive index of the optical disc to an incident light beam is rendered different if the material of the disc is different, so that the incident light quantity of the return light beam to the photo detector may also be changed. In addition, occurrence of an ambient temperature fluctuation brings about some harmful influences inclusive of a positional deviation of the objective lens due to the resultant temperature fluctuation in the apparatus.
Also, after the focus bias is determined by an experiment and the like, a determined focus bias is stored in a memory and the like which is included in the optical disc player. However, when the bias is thus pre-set, the focus bias is changed due to the aging of the optical disc player and thereby may incur the malfunction of the optical disc player.
Consequently, it become difficult to adjust the focus bias exactly to its optimal value in the reproduction of data from the optical disc.


Antianxiety drugs, also known as anxiolytic drugs or tranquilizers, are increasingly important psychotropic drugs. They include the benzodiazepines, such as diazepam, chlordiazepoxide, gidazepam, oxazepam, phenazepam, lorazepam, and the like. These benzodiazepines have been the most widely used drugs for the treatment of anxiety. Other antianxiety drugs include, buspirone, gepirone, ipsapirone, and other drugs with high affinity and selectivity for 5HT1.sub.a receptor sites. See for example, Abou-Gharbia et al., J. Med. Chem., 32, 1024 (1989).
These agents have been very effective but there has been increasing concern about the disadvantages associated with benzodiazepine therapy. The spectrum of their pharmacological activity, in addition to anxiolytic activity, includes sedative, anticonvulsant, miorelaxant, and amnestic effects, which are often considered both unnecessary and undesirable in the treatment of pathological anxiety. Thus a substantial need exists for nontoxic, highly active, selectively anxiolytic drugs without sedative, muscle relaxant and amnestic activities, which can be used for the treatment of anxiety. The present invention is directed to addressing this need.
A number of compounds having 1,2,3,4-tetrahydropyrrolo-[1,2-a]-pyrazine heterocycle have been reported. For example, nonsubstituted 1,2,3,4-tetrahydropyrrolo-[1,2-a]-pyrazine is described by A. Skoldinov et al. (U.S. Pat. No. 4,230,856 issued Oct. 28, 1980; USSR Patent No. 798,104 published Jan. 25, 1981). This compound is a precursor in the synthesis of octahydropyrrolo-[1,2-a]-pyrazine, which is useful for the synthesis of physiologically active drugs. 1- and 1,2-substituted 1,2,3,4-tetrahydropyrrolo-[1,2-a]-pyrazine derivatives, having antidepressant activity, are described by I. Jirkovsky (U.S. Pat. No. 4,188,389 issued Feb. 12, 1980). 1-substituted 1,2,3,4-tetrahydropyrrolo-[1,2-a]-pyrazines, having hypotensive activity, are described by V. Peresada et al., Khim-Farm. Zh., 21(9), 1054 (1987). 1-substituted 1,2,3,4-tetrahydropyrrolo-[1,2-a]-pyrazines, possessing coronary-dilating activity, are described by V. Peresada et al., Khim-Farm. Zh., 22(10), 1193 (1988).